Frequency multiplexed radio frequency identification

ABSTRACT

A radio frequency identification (RFID) system for frequency multiplexing includes, in an exemplary embodiment, an RFID interrogator configured for generating an RFID signal, wherein a channel frequency of the RFID signal changes over time within an operating bandwidth; and one or more electromagnetic transmissive elements each extending between a first end thereof and a second end thereof, each of the electromagnetic transmissive elements electrically coupled with the RFID interrogator at the first end thereof, each of the electromagnetic transmissive elements comprising a frequency dependent load at the second end thereof and configured for transmitting the RFID signal from the RFID interrogator to the frequency dependent load, wherein the frequency dependent load presents different electromagnetic characteristics to the RFID signal transmitted to the frequency dependent load depending on the channel frequency of the RFID signal. Other embodiments include a method for frequency multiplexing including similar components.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims priority to U.S. Provisional Application Ser.No. 62/356,121 titled “Systems, Apparatuses and Methods for FrequencyMultiplexed Radio Frequency Identification,” filed on Jun. 29, 2016, andis incorporated herein in its entirety by reference.

ORIGIN OF THE INVENTION

The invention described herein was made by employees of the UnitedStates government and may be manufactured and used by or for thegovernment of the United States of America for governmental purposeswithout the payment of any royalties thereon or therefor.

FIELD OF THE DISCLOSURE

The present disclosure relates generally to the field of radio frequencyidentification (“RFID”). More particularly, the disclosure relates tofrequency multiplexing in the context of RFID.

SUMMARY

Embodiments disclosed herein provide systems, methods, and apparatusesfor frequency multiplexed Radio Frequency Identification (RFID).

According to a first aspect of the disclosure, a radio frequencyidentification (RFID) system is provided. The system comprises an RFIDinterrogator configured for generating an RFID signal, wherein a channelfrequency of the RFID signal changes over time; a plurality of antennas;and a diplexer coupling the RFID interrogator and the plurality ofantennas and configured for distributing the RFID signal to each of theplurality of antennas, respectively, depending on the channel frequencyof the RFID signal generated. Each of the plurality of antennas isconfigured to transmit an electromagnetic wave in response to and at thechannel frequency of the RFID signal distributed thereto.

According to a second aspect of the disclosure, a radio frequencyidentification (RFID) system is provided. The system comprises an RFIDinterrogator configured for generating an RFID signal, wherein a channelfrequency of the RFID signal changes over time; a plurality oftransmission lines, each of the plurality of transmission lines being atleast partially open; and a diplexer coupling the RFID interrogator andthe plurality of transmission lines and configured for distributing theRFID signal to each of the plurality of transmission lines,respectively, depending on the channel frequency of the RFID signalgenerated. Each of the plurality of transmission lines is configured totransmit an electromagnetic signal in response to and at the channelfrequency of the RFID signal distributed thereto.

According to a third aspect of the disclosure, a radio frequencyidentification (RFID) system is provided. The system comprises an RFIDinterrogator configured for generating an RFID signal, wherein a channelfrequency of the RFID signal changes over time; a first narrow bandantenna, characterized by a first passband, the first passbandcorresponding to a first range of frequencies; a second narrow bandantenna, characterized by a second passband, the second passbandcorresponding to a second range of frequencies, wherein the second rangeof frequencies differs from the first range of frequencies, whereby thesecond passband differs from the first passband; and first and secondfeed lines configured for feeding the RFID signal to each of the firstand second narrow band antennas, respectively. The first narrow bandantenna is configured to transmit a first electromagnetic wave inresponse to and at the channel frequency of the RFID signal fed thereto,if the channel frequency of the RFID signal falls within the firstpassband, and the second narrow band antenna is configured to transmit asecond electromagnetic wave in response to and at the channel frequencyof the RFID signal fed thereto, if the channel frequency of the RFIDsignal falls within the second passband.

According to a fourth aspect of the disclosure, a radio frequencyidentification (RFID) system is provided. The system comprises an RFIDinterrogator configured for generating an RFID signal, wherein a channelfrequency of the RFID signal changes over time; at least one single feedpatch antenna; a first feed line configured for feeding the RFID signalto the single feed patch antenna, if the channel frequency of the RFIDsignal falls within a first passband; and a second feed line configuredfor feeding the RFID signal to the single feed patch antenna, if thechannel frequency of the RFID signal falls within a second passband, thesecond passband being different than the first passband. The single feedpatch antenna is configured to transmit a first electromagnetic wave inresponse to and at the channel frequency of the RFID signal fed theretofrom the first feed line and a second electromagnetic wave in responseto and at the channel frequency of the RFID signal fed thereto from thesecond feed line.

According to a fifth aspect of the disclosure, a radio frequencyidentification (RFID) system is provided. The system comprises an RFIDinterrogator configured for generating an RFID signal, wherein a channelfrequency of the RFID signal changes over time within a given bandwidth;at least one single feed patch antenna; and a single feed lineconfigured for feeding the RFID signal to the at least one single feedpatch antenna. The single feed patch antenna is configured to transmitan electromagnetic wave in response to and at the channel frequency ofthe RFID signal fed thereto from the feed line. The single feed patchantenna is further configured such that the electromagnetic waveexhibits (1) a polarization tilt angle that varies depending on thechannel frequency of the RFID signal, (2) a substantially linearpolarization at all channel frequencies of the RFID signal within thegiven operational bandwidth, and (3) a range of polarization tilt anglesacross the given operational bandwidth that spans at least 70 degreeswithin a single quadrant.

According to a sixth aspect of the disclosure, a radio frequencyidentification (RFID) system is provided. The system comprises an RFIDinterrogator configured for generating an RFID signal, wherein a channelfrequency of the RFID signal changes over time within an operatingbandwidth, the operating bandwidth comprising a first portion thereofand a second portion thereof, the first portion of the operatingbandwidth not completely overlapping with the second portion of theoperating bandwidth; at least one antenna; and a frequency selectivesurface. The RFID signal is to be fed to the least one antenna, and theleast one antenna is configured to transmit an electromagnetic wave inresponse to the RFID signal fed thereto. The frequency selective surfaceis configured to present as a boundary condition a surface impedancethat changes according to the frequency of an electromagnetic waveimpinging thereon. The frequency selective surface is configured suchthat (1) when the channel frequency of the RFID signal falls within thefirst portion of the operating bandwidth, a first surface impedance isestablished on the frequency selective surface that alters, according toa first pattern, the electromagnetic wave transmitted by the at leastone antenna and/or an electromagnetic wave transmitted by an RFID tagfor reception by the at least one antenna, and (2) when the channelfrequency of the RFID signal falls within the second portion of theoperating bandwidth, a second surface impedance is established on thefrequency selective surface that alters, according to a second pattern,the electromagnetic wave transmitted by the at least one antenna and/oran electromagnetic wave transmitted by an RFID tag for reception by theat least one antenna. The first pattern differs from the second pattern.

According to a seventh aspect of the disclosure, a radio frequencyidentification (RFID) system is provided. The system comprises an RFIDinterrogator configured for generating an RFID signal, wherein a channelfrequency of the RFID signal changes over time within an operatingbandwidth, the operating bandwidth comprising a first portion thereofand a second portion thereof, the first portion of the operatingbandwidth not overlapping with the second portion of the operatingbandwidth; and one or more electromagnetic transmissive elements eachextending between a first end thereof and a second end thereof each ofthe electromagnetic transmissive elements electrically coupled with theRFID interrogator at the first end thereof, each of the electromagnetictransmissive elements comprising a frequency dependent load at thesecond end thereof and configured for transmitting the RFID signal fromthe RFID interrogator to the frequency dependent load, wherein thefrequency dependent load presents different electromagneticcharacteristics to the RFID signal transmitted to the frequencydependent load depending on the channel frequency of the RFID signal.

According to an eighth aspect of the disclosure, a radio frequencyidentification (RFID) method is provided. The method comprisesgenerating an RFID signal, wherein a channel frequency of the RFIDsignal generated changes over time; distributing the RFID signal to aplurality of electromagnetic transmissive elements at different times,respectively, depending on the channel frequency of the RFID signalgenerated; transmitting a first electromagnetic signal having a firstchannel frequency in response to a first distributed RFID signal havingthe first channel frequency; and transmitting a second electromagneticsignal having a second channel frequency in response to a seconddistributed RFID signal having the second channel frequency. Theplurality of electromagnetic transmissive elements comprises a pluralityof antennas, a plurality of antenna feed lines, or a plurality of atleast partially open transmission lines.

According to a ninth aspect of the disclosure, a radio frequencyidentification (RFID) method is provided. The method comprisesgenerating an RFID signal, wherein a channel frequency of the RFIDsignal changes over time within a bandwidth; feeding the RFID signal toa microstrip antenna; and transmitting, by the microstrip antenna, anelectromagnetic wave in response to the RFID signal fed thereto, theelectromagnetic wave having a polarization that varies depending on thechannel frequency of the RFID signal fed thereto.

Other aspects and features of the embodiments described herein willbecome apparent from the following description and the accompanyingdrawings, illustrating the principles of the embodiments by way ofexample only.

BRIEF DESCRIPTION OF THE DRAWINGS

The following figures form part of the present specification and areincluded to further demonstrate certain aspects of the present claimedsubject matter, and should not be used to limit or define the presentclaimed subject matter. The present claimed subject matter may be betterunderstood by reference to one or more of these drawings in combinationwith the description of embodiments presented herein. Consequently, amore complete understanding of the present embodiments and furtherfeatures and advantages thereof may be acquired by referring to thefollowing description taken in conjunction with the accompanyingdrawings, in which like reference numerals may identify like elements,wherein:

FIG. 1 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID systemincluding a reader, a diplexer and a plurality of antennas.

FIG. 2 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID systemincluding a reader, a diplexer and a plurality of antennas, wherein thediplexer is implemented in a distributed fashion.

FIG. 3 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID systemincluding a reader, a diplexer and a plurality of antennas, wherein thesystem has a non-linear configuration.

FIG. 4 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID systemincluding a reader, a diplexer, a plurality of antennas, and abeamforming network.

FIG. 5 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID systemincluding a reader, a diplexer, a plurality of antennas, and a pluralityof beamforming networks, wherein the diplexer and the beamformingnetworks are implemented in a distributed fashion.

FIG. 6 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID systemfor a smart shelf, the system including a reader, a diplexer, and aplurality of at least partially open transmission lines, wherein thediplexer is implemented in a distributed fashion.

FIG. 7 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID systemfor a smart enclosure, the system including a reader and a plurality ofnarrow band antennas.

FIG. 8 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID narrowband antenna and feed system, including a reader and a plurality ofnarrow band antennas.

FIG. 9 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID narrowband antenna and feed system, including a reader and a plurality ofnarrow band antennas, the system providing polarization diversity.

FIG. 10 is a schematic diagram, in accordance with one or moreembodiments described herein, of a frequency multiplexing RFID narrowband antenna and feed system, including a reader and a single microstrippatch antenna fed by two feed lines, the system providing polarizationdiversity.

FIGS. 11A and 11B are schematic diagrams, in accordance with one or moreembodiments described herein, each illustrating a respective frequencymultiplexing RFID narrow band antenna and feed system, including areader and a single microstrip patch antenna fed by a single feed line,the system providing polarization diversity.

FIG. 12 is a schematic diagram, in accordance with one or moreembodiments described herein, of determination of RFID tag orientationbased on polarization, which may be performed by a frequencymultiplexing RFID system.

FIG. 13 is a schematic diagram, in accordance with one or moreembodiments described herein, of an arrangement employing a frequencyselective surface in the context of frequency multiplexing RFID.

FIG. 14 is a schematic diagram, in accordance with one or moreembodiments described herein, of an arrangement employing a frequencyselective surface in the context of frequency multiplexing RFID, showingthe effects of the frequency selective surface on electric fields ofelectromagnetic waves used in the frequency multiplexing RFID.

FIG. 15 is a schematic diagram, in accordance with one or moreembodiments described herein, illustrating a layered structure of afrequency selective surface.

FIG. 16 is a schematic diagram, in accordance with one or moreembodiments described herein, of an arrangement employing a frequencyselective surface and a current sheet in the context of frequencymultiplexing RFID.

FIG. 17 is a schematic diagram, in accordance with one or moreembodiments described herein, of an arrangement employing a frequencydependent load in the context of frequency multiplexing RFID.

FIG. 18 is a flow chart, in accordance with one or more embodimentsdescribed herein, of a frequency multiplexing RFID method.

FIG. 19 is a flow chart, in accordance with one or more embodimentsdescribed herein, of a frequency multiplexing RFID method.

FIG. 20 is a graph showing the normalized magnitude voltage responses ofupper and lower resonant modes in a single feed circularly polarizedpatch antenna.

FIG. 21 is a graph showing the normalized magnitude voltage responses ofupper and lower resonant modes in a single feed circularly polarizedpatch antenna having a higher unloaded quality factor as compared toFIG. 20.

FIG. 22 is a graph showing the normalized magnitude voltage responses ofupper and lower resonant modes in a single feed multi-linear polarizedpatch antenna, in accordance with one or more embodiments describedherein.

FIG. 23 is a graph showing the (A) polarization vectors, (B)polarization tilt angles, and (C) total angular span covered, (i) forthe case of a single feed circularly polarized patch antenna and (ii)for the case of a single feed multi-linear polarized patch antenna, inaccordance with one or more embodiments described herein.

NOTATION AND NOMENCLATURE

Certain terms are used throughout the following description and claimsto refer to particular system components and configurations. As oneskilled in the art will appreciate, the same component may be referredto by different names. This document does not intend to distinguishbetween components that differ in name but not function. In thefollowing discussion and in the claims, the terms “including” and“comprising” (and the like) are used in an open-ended fashion, and thusshould be interpreted to mean “including, but not limited to. . . .”Also, the term “couple”, “coupled” or “couples” (and the like) isintended to mean either an indirect or direct connection. Thus, if afirst device couples to a second device, that connection may be througha direct connection, or through an indirect connection via other devicesand connections. The word “or” is used in the inclusive sense (i.e.,“and/or”) unless a specific use to the contrary is explicitly stated.The word “substantially” (where not already preceded by the words “atleast”) shall be construed to mean “at least substantially.”

It should be noted that the terms “radio frequency” (RF) and “microwave”are used interchangeably herein. The terms “interrogator” and “reader”are likewise used interchangeably to connote a transceiver thattransmits electromagnetic radiation to one or more RFID tags andreceives responses from the one or more RFID tags. While theinterrogator may be operationally coupled to one or more processors,such processors may be internal and/or external to the interrogator. Forexample, in some cases the interrogator may have an internal or embeddedprocessor that controls the functionality of the interrogator and isalso capable of decoding and utilizing information received from one ormore tags. In other cases, the interrogator might have an internal orembedded processor that controls the communication functionality of theinterrogator, and an interface to an external processor enables theexternal processor to utilize information received from the one or moretags.

Although there is not unanimous concurrence regarding the definition of“waveguides” and “transmission lines,” the consensus opinion is thattransmission lines are a subset of waveguides that propagate,predominantly, transverse electromagnetic (TEM) waves. Herein, the term“transmission line” is used in a more general sense to denote anelongated device for transferring electromagnetic energy between twopieces of equipment, regardless of the specific propagation modesestablished within the elongated device. Although the term “waveguide”sometimes is construed to mean a hollow elongated, usually conductive,tube, the intent in this detailed description is the more generalmeaning relating to any structure designed to propagate anelectromagnetic field in one or more intended directions.

The terms “pattern,” “antenna pattern,” “(antenna) radiationdistribution pattern” or the like used herein pertain to the radiationdistribution produced over a solid angular region by an antenna inresponse to an injection of electromagnetic energy within a specificoperating frequency band or set of operating frequency bands. Thepattern may comprise one or more primary beams, wherein “beam” is usedto denote a pattern of radiation density over an angular span thatcontains a peak radiation density, and “beam” can also be described as amajor lobe. In addition to the radiation intensity produced as afunction of angle, “pattern” may also convey the variation inpolarization as a function of angle. It is possible for a pattern tocontain multiple lobes or beams, each lobe or beam characterized by alocal maximum of radiation density. (It will be understood that the word“pattern” by itself may also be used herein to refer not to an antennaradiation distribution pattern but rather to another kind of pattern,whether of radiation or not. Context will make clear the meaning of theword “pattern” when used by itself; for example, a pattern that is notan antenna radiation distribution pattern will not be presented asinherently associated with a particular antenna.)

The term “polarization vector” is used herein to describe thepredominant polarization exhibited by an antenna at a particularfrequency; that is, it is used synonymously with the major axis of anelliptically polarized electromagnetic field. Although an ellipticallypolarized field also exhibits instantaneous electromagnetic fieldscharacterized by polarization vectors not aligned with the major axes,the term “polarization vector” is used herein to convey the polarizationdefined by the major axis of the polarization ellipse, unless explicitlystated otherwise or made apparent by the context of the description.

The term “localize” (and the like) refers to identifying the location orposition of an item, either in a global fixed coordinate system orrelative to some other item or coordinate system. Identifying thelocation or position of an item implies also detecting the presence orexistence of an item. In addition, identifying the position of an itemover time may effectively serve to identify its distance (e.g., distancebetween RFID tag (attached to an item) and reader or other fixed object)and its movement (e.g., speed, direction, etc.) and to back an item overtime. Relatedly, the term “ranging” (and the like) refers to determiningthe distance of a tag from a reader (or other location), or vice versa.Again, distance may be deemed to imply location, presence, movementcharacteristics, etc. This disclosure also discusses determining theorientation or bearing of an item, which is related to but distinct fromlocalization.

As used herein, the term “frequency multiplexing” (and the like) refersto an arrangement in which a signal is selectively distributed (or fedor transmitted) to one of multiple electromagnetic transmissive elements(e.g., one of multiple antennas, one of multiple feed lines, one ofmultiple transmission lines, etc.), depending on the frequency of thesignal, and/or an arrangement in which signals excite different modes orpolarizations, depending on the frequency of the signal. Sucharrangements can also be combined. Frequency multiplexing may operateaccording to a first manner of operation such that when the frequency ofa signal (e.g., generated by an RFID reader) is within a first band thesignal is distributed (or fed or transmitted) to a first antenna (ortransmission line, feed line, etc.), when the frequency of the signal iswithin a second band the signal is distributed (or fed or transmitted)to a second antenna (or transmission line, feed line, etc.), and so on.Frequency multiplexing may operate according to a second manner ofoperation such that when the frequency of a signal (e.g., fed to anantenna, etc.) is within a first band the antenna transmits a signalhaving a first polarization, when the frequency of the signal is withina second band the antenna (or a different antenna) transmits a signalhaving a second polarization, and so on, the first and second (and anyadditional) polarizations having respective different orientations.Thus, the second manner of operation may be implemented in a systemhaving only one antenna or in a system having multiple antennas. In sum,frequency multiplexing serves effectively to switch or route a signalbetween different antennas (or between different transmission lines,between different feed lines, etc.) and/or between differentpolarizations or modes. Generating or transmitting a signal having acertain polarization may also be referred to as exciting a certain modeof a device such as an antenna, waveguide, or cavity. It will beunderstood that this switching or routing between different antennas,polarizations, etc. occurs over time. One of ordinary skill in the art,now having the benefit of this disclosure, will appreciate that, in thiscontext, reference to frequency of a signal in application refers to thecenter frequency or carrier frequency of a modulated signal and not tothe instantaneous frequency of a signal, which is constantly changing.In this disclosure, the application or use of frequency multiplexing inthe context of RFID may be referred to as “frequency multiplexing RFID”or “frequency multiplexed RFID” interchangeably. In this disclosure, theterms “frequency multiplexing” and “frequency multiplexed” may beabbreviated as “FM.”

“Frequency multiplexing” as the term is used herein is distinct from anumber of similar terms. For example, frequency multiplexing is not thesame as frequency-division multiplexing, which is a technique by whichthe total bandwidth available in a communication medium is divided intoa series of non-overlapping frequency sub-bands, each of which is usedto carry a separate signal, allowing a single transmission medium suchas the radio spectrum, a cable, or optical fiber to be shared bymultiple separate signals. Again, frequency multiplexing is not the sameas mere multiplexing, as that term is used in various contexts to meanswitching using a switch. It will be noted that frequency multiplexingmay be achieved by passive means (e.g., a filter) and does not requireor involve a switch (e.g., electrical, mechanical, etc.), activecontrols, a controller (whether machine or human), a power supply, etc.

The term “electromagnetic transmissive elements” is used herein to referto elements that guide, direct or channel electromagnetic energy, andthe term includes, e.g., antennas, antenna feed lines, microstrip lines,transmission lines, waveguides, etc. The terms “guide, direct orchannel” are not intended to be limited to an antenna or radiativefunctionality and are intended to be broader than and to encompass theterms “transmit” and “receive.”

With regard to systems, apparatuses and methods for frequencymultiplexed radio frequency identification described herein, the term“operating bandwidth” (or “operational bandwidth”) refers to thecumulative bandwidth of all channels (all frequencies) being used in thesystem, apparatus, or method (which cumulative bandwidth may also bereferred to as the “operating spectrum”). In some RFID systems, theoperating bandwidth may be divided into a number of narrowerband-limited channels, and the interrogator operates within one of thesechannels while communicating with RFID tags. The interrogator maysuccessively hop through these defined channels of the system(“frequency hopping,” discussed below). In passive tag RFID, the tagresponse is often offset slightly in frequency from the interrogatortransmit frequency to prevent it from being obscured. As used herein,the term “channel” will imply the band that includes both theinterrogator and the tag response. The term “channel frequency” willimply the narrow bandwidth of frequencies that define the channel, andthis term will be used in place of the terms “center frequency” and“carrier frequency, which were discussed above. As will be clear fromcontext, the operating (or operational) bandwidth may sometimes bereferred to simply as the “bandwidth.” In some cases, the bandwidth of aparticular antenna or other component, rather than of a systemcomprising a plurality of antennas or components, may be discussed.

With regard to systems, apparatuses and methods for frequencymultiplexed radio frequency identification described herein, lengths ofportions of waveguides, transmission lines, feed lines, etc. are oftendescribed in terms of (multiples of fractions of) a wavelength. Unlessindicated to the contrary, such wavelength refers to the wavelengthcorresponding to the frequency at the center of the operating bandwidth(and such wavelength may also be referred to as the wavelength of thecenter of the operating bandwidth).

DETAILED DESCRIPTION

The foregoing description of the figures is provided for the convenienceof the reader. It should be understood, however, that the embodimentsare not limited to the precise arrangements and configurations shown inthe figures. Also, the figures are not necessarily drawn to scale, andcertain features may be shown exaggerated in scale or in generalized orschematic form, in the interest of clarity and conciseness. Relatedly,certain features may be omitted in certain figures, and this omissionmay not be explicitly noted in all cases.

While various embodiments are described herein, it should be appreciatedthat the embodiments described herein encompass many inventive conceptsthat may be embodied in a wide variety of contexts. The followingdetailed description of exemplary embodiments, read in conjunction withthe accompanying drawings, is merely illustrative and is not to be takenas limiting the scope of the claims, as it would be impossible orimpractical to include all of the possible embodiments and contexts inthis disclosure. Upon reading this disclosure, many alternativeembodiments will be apparent to persons of ordinary skill in the art.The scope of the invention is defined by the appended claims andequivalents thereof.

Illustrative embodiments are described below. In the interest ofclarity, not all features of an implementation of the exemplaryembodiments may be described or illustrated in this specification. Inthe development of any such embodiment, numerous implementation-specificdecisions may be made to achieve the design-specific goals, which mayvary from one implementation to another. It will be appreciated thatsuch a development effort, while possibly complex and time-consuming,would nevertheless be a routine undertaking for persons of ordinaryskill in the art having the benefit of this disclosure.

RFID technologies may be categorized into “sparse zone” and “dense zone”technologies. The term “dense zone” is used to refer to smart enclosuresor shelves in which the electromagnetic fields are to some extentcontained. The term “sparse zones” is used to refer to those areasoutside of the dense zones, hence including generally open areas, butalso cracks, crevices, or other obscured spaces that are not covered bydense zone technologies. It should be noted that it is possible to haveoverlap between dense and sparse zones, especially in those cases inwhich the dense zones are not fully shielded. For example, in the caseof so-called “smart shelves,” items on the shelf might be read byembedded transmission lines, but they might also be read by a zonereader in the general vicinity. Embodiments disclosed herein pertain toboth dense and sparse zones, including those regions in which the zonesoverlap.

The most prevalent UHF RFID system worldwide is the EPCglobal Class 1Generation 2 protocol (synonymous with ISO 18000-6C), which describesthe communications between the interrogator, or reader, and the tag.This system enables efficient communications between a single reader andthousands of tags, and it is characterized by a narrow bandwidth. Whilehigh gain antennas can be used to achieve fine spatial resolution, suchhigh gain antennas also generally require, at UHF frequencies, largeantenna apertures. Such large apertures may be impractical withinconfined environments such as space habitats, e.g., the InternationalSpace Station (ISS).

Various embodiments will now be described. Embodiments disclosed hereinemploy systems and methods that may be referred to as frequencymultiplexing, frequency multiplexed RFID, frequency multiplexedlocalization, or the like.

As discussed above, embodiments disclosed herein teach arrangements inwhich a signal is selectively distributed (or fed or transmitted) to oneof multiple electromagnetic transmissive elements (e.g., one of multipleantennas, one of multiple feed lines, or one of multiple transmissionlines), depending on the channel frequency of the signal, andarrangements in which signals excite different modes or polarizations,depending on the channel frequency of the signal. Such arrangements canalso be combined. Thus, according to a first manner of operation,frequency multiplexing RFID may operate such that when the channelfrequency of a signal (e.g., generated by an RFID reader) is within afirst band the signal is distributed (or fed or transmitted) to a firstantenna (or transmission line, feed line, etc.), when the channelfrequency of the signal is within a second band the signal isdistributed (or fed or transmitted) to a second antenna (or transmissionline, feed line, etc.), and so on. According to a second manner ofoperation, frequency multiplexing may operate such that when the channelfrequency of a signal (e.g., fed to an antenna, etc.) is within a firstband the antenna transmits a signal having a first polarization, whenthe channel frequency of the signal is within a second band the antenna(or a different antenna) transmits a signal having a secondpolarization, and so on, the first and second (and any additional)polarizations (i.e., polarization vectors) being characterized byrespective different orientations. Thus, the second manner of operationmay be implemented in a system having only one antenna or in a systemhaving multiple antennas. In sum, frequency multiplexing serveseffectively to switch or route a signal between different antennas (orbetween different transmission lines, between different feed lines,etc.) and/or between different polarizations or modes.

According to a first described (set of) embodiment(s) (described belowwith reference to FIG. 1), RFID signals of different channel frequenciesmay be transmitted to respective different regions, and the location ofa tagged item may be determined by reference to the channel frequency ofthe response signal sent by the item's RFID tag in conjunction with thecoverage area associated with that channel frequency.

FIG. 1 is a schematic diagram of a frequency multiplexing (FM) RFIDsystem 100 including an RFID reader or interrogator 105, a diplexer 110,and a plurality of reader antennas 115.

Reader 105 has the capability to broadcast over a range of channels,each channel corresponding to a different frequency band. RFIDcommunication protocols typically utilize very narrow bands or channelsover an allowed range of the frequency spectrum governed by a regionalregulatory authority. In the United States, for example, the EPCglobalClass 1 Generation 2 protocol is required by the Federal CommunicationsCommission (FCC) to implement Frequency Hopping Spread Spectrum (FHSS)over the range 902-928 MHz. FHSS is a method of transmitting radiosignals by rapidly switching among many frequency channels, using apseudorandom sequence. Each channel is typically very narrow, oftenabout 500 kHz. Reader 105 may employ an FHSS approach, which alternatesbetween different channels over time, i.e., generating a signal at achannel frequency within a first bandwidth for a first duration of time,then generating a signal at a channel frequency within a secondbandwidth for a second duration of time, and so on (as mentioned, thesequence of alteration may be pseudorandom; channels may be repeatedover time). According to embodiments disclosed herein, FM RFID mayemploy channels (bandwidths) as narrow as those utilized by theEPCglobal Class 1 Generation 2 protocol, or channels (bandwidths)covering a broader range of the spectrum. FM RFID may be designed tosupport a multitude of the FHSS channels.

FM RFID system 100 further includes a multi-channel diplexer 110,disposed between reader 105 and antennas 115, to distribute the signalgenerated by reader 105 to the antennas 115, as shown schematically(channels 1-4) in FIG. 1. Diplexer may comprise, e.g., a plurality offilters, such as bandpass filters, such that the signal generated byreader 105 is distributed to a respective one of the antennas 115depending on the channel frequency of the signal. For example, withreference to FIG. 1, a first bandpass filter may pass only signalshaving a frequency within a first bandwidth (channel 1, say, 902-903MHz) to a first (leftmost, in FIG. 1) antenna 115, a second bandpassfilter may pass only signals having a frequency within a secondbandwidth (channel 2, say, 904-905 MHz) to a second (left middle, inFIG. 1) antenna 115, a third bandpass filter may pass only signalshaving a frequency within a third bandwidth (channel 3, say, 906-907MHz) to a third (right middle, in FIG. 1) antenna 115, and a fourthbandpass filter may pass only signals having a frequency within a fourthbandwidth (channel 4, say, 908-909 MHz) to a fourth (rightmost, inFIG. 1) antenna 115. While this example employs bandpass filters, it isalso possible to use lowpass and highpass filters. While this exampleemploys non-overlapping channels, it is also possible to employoverlapping channels. With use of diplexer 110, the use of a switch todistribute the signal of interrogator 105 is eliminated. The diplexer110 may be a passive device (e.g., a filter such as a surface acousticwave filter). As such, diplexer 110 does not require power, cabling,control signals, or control logic, as is used for a conventionaldistributed switched antenna system.

In response to the signal distributed to an antenna 115 (any one of theantennas 115) by diplexer 110, the antenna 115 transmits a signal(electromagnetic wave) having the channel frequency of the signaldistributed thereto by diplexer 110. An RFID tagged item within therange of the transmitted electromagnetic wave transmits a signal(electromagnetic wave) in response to the transmitted electromagneticwave. The response signal is received (most strongly) by one of theantennas 115, and sent to reader 105. Reader 105, e.g., in conjunctionwith associated processing logic, infers the location (or presence,distance, orientation, etc.) of the RFID tagged item) based on thechannel frequency (channel) of the response signal and the antennacoverage area associated with that channel (the coverage area of the oneof the antennas 115 that transmits and receives on that channel). Thatis, the RFID tagged item is determined to be in the coverage area of theparticular one of the antennas 115 that transmits and receives on thechannel used by the response signal. In the example shown in FIG. 1, theRFID tagged item, identified as “ID3,” is determined to be located inthe coverage area of the fourth (rightmost) antenna 115 because theresponse signal received from that RFID tag was received (most strongly)at channel 4. Thus, as reflected by the fact that the coverage areas aredesignated as Ch1-Ch4 in FIG. 1, in system 100 the respective coverageareas of antennas 115 correspond to the respective channels 1-4.

With further regard to the determination of location of the RFID tag,the operation thereof may be as follows. The response signal from thetag may be received by a single one of antennas 115 or by multipleantennas 115 (as discussed, depending on the channel frequency of theresponse signal). If the response signal is received by only a singleantenna 115, the location of the tag would correspond to the coveragearea of that antenna. If the response signal is received by multipleantennas 115, the received signal strengths on all receiving paths(receiving antennas 115) would be compared. The tag location estimatewould be weighted according to the received signal strengths (asdiscussed, the received signal strengths collectively depend on thechannel frequency of the response signal).

The determination of location of the RFID tag may also operate asfollows. The (electromagnetic) spectral response of the tag mightindicate a particular location with a non-obvious association. Forexample, in a complex scattering environment, a given location of a tagmay result in specific recognizable spectral responses in each of two ormore different antennas 115 due to the propagation channels establishedby each antenna 115 at each channel frequency. The system may learn theassociation between a given set of spectral responses received by agiven set of antennas 115 and a given tag location. Such associationsmay be learned for multiple tag locations.

The remarks given above regarding determination of location in system100 may apply to embodiments throughout this disclosure generally, notjust to system 100. Also, although these remarks refer to determinationof “location,” they are intended to cover the full range of associatedinformation (such as distance, presence, orientation/bearing, speed,direction, tracking information, etc.), as discussed elsewhere in thisdisclosure. The term “state” of an item may be used to refer to all or aportion of this information (i.e., location and associated information).

It should be noted that, for non-RFID systems, frequency multiplexing(FM) is typically achieved using pre-designed signal frequencies thatare well separated so that filter bands are fairly distinct withsufficient separation such that the filter isolation between the twobands can be made very high. This separation is used because, in suchother FM systems, transmission in any of the bands might occursimultaneously, so isolation between the bands help avoid interference.In such other FM systems, a given signal is fully in one band or anotherwith considerable isolation between the bands, and there is a one-to-onecorrespondence between the defined physical operating bands (that wouldbe defined by filters or diplexers) and the channels on which thetransceivers operate. In addition, in such FM systems, different signalsoften flow simultaneously, thus necessitating the clear separation thatis provided by having each signal contained in a band that is isolatedfrom other bands. For example, multiple radio clients might each beserviced simultaneously by a distinct FM band, and isolation will avoidinterference.

In contrast, in the context of FM RFID, the channels are generally veryclosely spaced and there are more distinct channels. The channel spacingis particularly close and the number of channels particularly high whenthe primary intent of the channelization is for FHSS. As describedherein, only one channel is used at any given time so that isolationbetween adjacent channels is not much of a concern. The RFID FM systemsdescribed herein may have many more channels defined than segmentationsdefined by filters, diplexers, or other frequency dependent devices.Thus, some channel content might flow through more than one of saidsegmentations. The channelization scheme of this type of RFID thus wouldgenerally not be deemed well designed or well suited for other types ofFM, because the channels are too closely spaced and hence would requirefilters that could not be realized in practice without unacceptable lossor size. However, FM is successfully applied in the RFID context asdescribed in the instant disclosure by virtue of the highly statisticalnature of RFID links, and, in some instances, application of artificialintelligence to derive conclusions or statistical inferences. Forexample, in many RFID scenarios, the reader interrogates the same tagmany hundreds or thousands of times within a minute. Due to the closelyspaced RFID channel bands, the RFID FM devices might not fully divertall power in its entirety as they would in other FM devices, so thepower is smeared to more than one device (e.g., antenna, filter,waveguide). Said another way, the directivity of the FM devices in RFIDapplications is likely not as high as in other FM applications. As anexample, with FM applied to an RFID application, channel frequencies ata low end of the operating spectrum might be directed strongly toward afirst antenna direction or polarization. Channel frequencies at theother (i.e., high) end of the operating spectrum would be directedstrongly toward a second antenna direction or polarization. Channels inbetween might direct some portion of power toward the first antennadirection or polarization with the remainder of power being directedtoward the second antenna direction or polarization, with the ratiotransitioning from predominance toward the first to predominance towardthe second as the channel frequency increases. So while thecharacteristics of the communication channel owing to the interrogatorchannel frequency might not be conducive to a strong reader-tag link atany given time interval, over many trials in different channels, withthe communication channel undergoing modifications from one trial to thenext due to a frequency multiplexed influence, the system is more likelyto succeed as compared to using other RFID techniques. Moreover, ifallowed by regional governing regulations, the system can learn which FMconfigurations (channels) are optimal for communicating with each tag.This type of intermittent or variable performance would likely not betolerable in other communication applications, for example, cellularcommunications. In some applications, the technology described hereinare practical because of the inherent RFID range limitations imposed bysafety and regulatory bodies. Diplexers can often be made verynarrowband at the expense of efficiency, to the point where thedecreased efficiency renders the diplexer impractical for othercommunication links if the channels were as narrow as in many RFIDsystems. However, antenna gain, transmit power, or both are oftenlimited by regional regulations or by safety considerations. So, thepower inefficiency of a narrowband diplexer can be inconsequential inmany RFID links, and in some embodiments described herein, FM applied toRFID can result in improved performance that would not be possible byincreasing transmit power or antenna gain due to regulatory or safetyconstraints. In other embodiments described herein, antenna designs areemployed in novel ways to achieve integrated narrowband multiplexingbased on the inherent band-limited features of these particular antennadesigns, which heretofore have been considered only as limitations. Theremarks in this paragraph apply generally to embodiments describedherein.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 2. FIG. 2 is a schematic diagram of an FM RFID system200 including a reader 205, a diplexer 210 and a plurality of readerantennas 215, wherein the diplexer 210 is implemented in a distributedfashion. In the arrangement shown in FIG. 2, diplexer 210 includes awaveguide 220, a plurality of (e.g., quarter-wave) impedancetransformers 225, and a plurality of filters 230. Each impedancetransformer 225 is coupled to a respective filter 230, and each filter230 is coupled to a respective antenna 215. As described with referenceto FIG. 1, each filter 230 may be a bandpass filter that permits asubset of the total channel spectrum to pass between the reader 205 andthe respective antenna 215. The out-of-band impedance looking into thefilters 230 may closely approximate a short-circuit, so distributedλ/4-wave impedance transformers 225 are connected between respectivefilters 230 and waveguide 220 such that the impedance at waveguide 220looking into any of the transformers 225 approximates an open-circuit atout-of-band frequencies. Here and in further discussions throughout thisdisclosure pertaining to wavelength and the lengths of transmissionlines, λ is the wavelength at the center of the operating bandwidth,i.e., the cumulative bandwidth of all channels being used (i.e., in FIG.2, channels 1-4) (λ may also be referred to as the effectivewavelength), unless indicated to the contrary. Similarly, each interface235 between a respective antenna 215 branch and waveguide 220 isseparated from an adjacent interface 235 by an integer number of½-wavelengths along waveguide 220 such that any open circuit at anantenna 215 branch is transformed to appear as an open circuit at any ofthe other interfaces 235. (In FIG. 2, m₁, m₂ and m₃ represent integers,as do n₁ and n₂, and L equals ½ wavelength.)

The arrangement of diplexer 210 in FIG. 2 described above may also bedescribed as follows. Diplexer 210 includes waveguide 220 and aplurality of quarter-wave transformers 225, each of the quarter-wavetransformers 225 coupling the waveguide 220 with a respective one of thefilters 230, which interface respectively to the antennas 215, each ofthe quarter-wave transformers 225 forming a respective interface (i.e.,junction point) 235 with the waveguide 220. Reference numerals 222represent merely connections between quarter-wave transformers 225 andrespective filters 230; they do not represent physical lengths of feedlines between the quarter-wave transformers 225 and the filters 230.Diplexer 210 further includes a plurality of bandpass filters 230, eachof the bandpass filters 230 disposed between and coupled to a respectivewave transformer 225 and a respective antenna 215. Each of the bandpassfilters 230 is characterized by a passband and an impedance that issubstantially zero at frequencies outside of the passband. Adjacent onesof the interfaces 235 are separated by a respective portion of thewaveguide 220 having a length equal to a respective integer (m) multipleof a half of a wavelength, the wavelength being that wavelengthcorresponding to the center of the system operating bandwidth, thesystem operating bandwidth comprising the span of frequencies of allchannels employed in the RFID system. The RFID interrogator 205 isconnected to the waveguide 220 at a junction 207, a distance between thejunction 207 and either of the two closest interfaces 235 being aninteger (n) multiple of one half of a wavelength. Each of thequarter-wave transformers 225 may be a quarter-wave impedancetransformer. It is noted that the quarter-wave transformers 225 may havea length equal to any odd integer of a quarter-wavelength.Alternatively, it is noted that the quarter-wave transformers 225 mayhave a length equal to any even integer of a quarter-wavelength if eachof the bandpass filters 230 is characterized by a passband and animpedance that is very large relative to the impedances of connectingdevices 215 and 225 at frequencies outside of the passband.

In some embodiments, the channel filters 230 are implemented usingsurface acoustic wave (SAW) technology, as SAW filters can beimplemented with very low loss for very narrow bandwidths. It should benoted that the channels designated as Ch1-Ch4 do not necessarilycorrespond to the channels of a particular RFID standard or protocol,nor are they necessarily of the same bandwidth. For example, thebandwidth Ch1 may correspond to 3 channels of the UHF standard EPCglobalClass 1 Generation 2, whereas the bandwidth Ch2 may correspond to 5channels of the same UHF standard. In some embodiments, the passbandsassociated with one or more channels are not contiguous. Although thecoverage areas corresponding with each channel are shown asnon-overlapping, in some embodiments the system is designed such thatthese coverage areas overlap. The remarks in this paragraph apply notonly to the embodiments described here with reference to FIG. 2, butalso to the embodiments described above with reference to FIG. 1 and theembodiments described below with reference to subsequent drawings.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 3. FIG. 3 is a schematic diagram of an FM RFID system300 including a reader 305, a diplexer 310 and a plurality of readerantennas 315, where the system has a non-linear configuration. Thenon-linear configuration may be deemed a rotated, circular or partialcircular configuration. The arrangement shown in FIG. 3 includeselements analogous to those of FIG. 2 (e.g., transmission line 320,quarter-wave transformers 325, filters 330, and interfaces 335), but thecharacteristics of those elements or of the overall system (beyond thenon-linear configuration) are not necessarily identical to that of FIG.2. As with system 200 of FIG. 2, analogously in system 300 of FIG. 3,diplexer 310 includes the waveguide 320, at least one of thequarter-wave transformers 325 and at least one of the filters 330, asindicated by the bracket shown in FIG. 3. (Note that the lines shownbetween respective quarter-wave transformers 325 and respective filters330, analogous to elements 222 in FIG. 2, represent mere connections,not physical lengths of feed/transmission line.) In system 300, antennas315 are arranged in a rotated (non-linear) configuration (rather thanthe linear configuration of system 200) so as to cover a wider angle,provide higher gain than a simple wide-beam or omni-directional antenna,and permit localization of a tag based on the frequencies at which tagsrespond. Transmission line (waveguide) 320, wave transformers 325,filters 330, and interfaces 335 are also arranged in a rotated orcircular configuration corresponding to that of antennas 315. Theangular separation between antennas 315 and the entire angular extent ofthe rotated configuration of antennas 315 in system 300 may be otherthan that shown in FIG. 3. In system 300, restrictions similar oridentical to those discussed with respect to system 200 regarding theseparation of interfaces and the electrical length of the transmissionlines may apply. Alternatively, the configuration of system 300 may becompact in size so that all of the interfaces 335 are electrically closeto each other, thus eliminating the requirement that the spacing betweenthe interfaces be integer numbers of half-wavelengths.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 4. FIG. 4 is a schematic diagram of an FM RFID system400 including a reader 405, a diplexer 410, a plurality of readerantennas 415, and a beamforming network 440. Beamforming network 440 isdisposed between diplexer 410 and antennas 415. (It will be understoodthat the language “disposed between” does not necessarily mean thatbeamforming network 440 is disposed physically/spatially betweendiplexer 410 and antennas 415, but rather that beamforming network 440is disposed operationally between diplexer 410 and antennas 415, thatis, signals are transmitted from diplexer 410 to antennas 415 viabeamforming network 440.) Beamforming network 440 includes terminalports 445 and antenna ports 450. The signals to and from reader 405 arechanneled by diplexer 410 to ones of the terminal ports 445 ofbeamforming network 440. The antenna ports 450 of beamforming network440 are each connected to a respective antenna 415. As those of ordinaryskill in the art would understand, now having the benefit of thisdisclosure, beamforming network 440 and antennas 415 act to form acharacteristic set of antenna radiation distribution patterns 460, witheach radiation distribution pattern 460 associated with one of thebeamforming network terminal ports 445. Thus, each of the four channels1-4 designated on diplexer 410 is associated with one of the fourradiation distribution patterns 460 as shown. (Such a set of radiationdistribution patterns may conventionally be shown relative to aCartesian coordinate system defined by x and y axes and an origin O, asshown.) The approach illustrated in FIG. 4 may permit long rangeinterrogation, as the signals received by the antennas are combined, andsimilarly, signals transmitted by the antennas are combined, to form astrong, composite signal.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 5. FIG. 5 is a schematic diagram of an FM RFID system500 including a reader 505, a diplexer 510, pluralities 515, 516 ofreader antennas, and a plurality of beamforming networks 540, 541,wherein the diplexer 510 (subdiplexers 510 a and 510 b) and thebeamforming networks 540, 541 are implemented in a distributed fashionsimilar to that shown in FIG. 2. System 500 (FIG. 5) may thus be thoughtof as an extension of system 400 of FIG. 4, in which multiplebeamforming networks 540, 541 and a diplexer are distributed in a mannersimilar to system 200 (as shown in FIG. 2). Thus, system 500 (FIG. 5)may be understood conceptually as a combination of (the definingfeatures of) system 200 (FIG. 2) and system 400 (FIG. 4). In system 500,the spacing between interfaces, whether between adjacent diplexedbeamforming networks 540, 541 or between a diplexed beamforming network540 or 541 and reader 505, may be constrained as discussed inconjunction with system 200 (FIG. 2) such that out-of-band circuits donot result in impedance mismatches at other interfaces. For example, itis assumed that the subdiplexers 510 a and 501 b include filters andimpedance transformers (analogously to systems 200 and 300) to isolatechannels. As described with reference to system 400 (FIG. 4),beamforming networks 540, 541 and antennas 515, 516 act to formcharacteristic sets of antenna radiation distribution patterns 560, 561.Although shown with only two beamforming networks 540, 541, system 500is extensible to a greater number of beamforming networks. Furthermore,although the antennas 515, 516 of the beamforming networks 540, 541 areshown in FIG. 5 to be spaced in a collinear fashion, the system isextensible to rotated (non-linear) configurations in which each array517, 518 of antennas 515, 516 is rotated in a circular/partly circularconfiguration similar to the rotation of single antenna elements 315 insystem 300 shown in FIG. 3. That is, e.g., array 517 of antennas 515 maybe rotated relative to array 518 of antennas 516, with the antennas 515arranged linearly relative to one another and the antennas 516 arrangedlinearly relative to one another. In this case, each of arrays 517, 518may be referred to as a linear array. However, it is also possible forthe antennas 515 to be arranged in a rotated configuration relative toone another, and/or the antennas 516 to be arranged in a rotatedconfiguration relative to one another. In this case, each of arrays 517,518 may be referred to as a rotated array. Diplexer 510 may beunderstood as encompassing sub-diplexer 510 a for beamforming network540, sub-diplexer 510 b for beamforming network 541, andwaveguide/transmission line 520 connecting reader 505 with sub-diplexer510 a and sub-diplexer 510 b. In the context of FIG. 5, each ofsub-diplexers 510 a and 510 b presents an open-load impedance (i.e.,effectively infinite) over passband channels of the other ofsubdiplexers 510 a and 510 b, and the spacings between the readerjunction 507 at waveguide/transmission line 520 and the interfaces 535(of sub-diplexers 510 a and 510 b) at waveguide/transmission line 520are such that this open load-impedance is presented to the reader 505 onthe side for which the channel frequency is not within the passband ofthat respective sub-diplexer. In another embodiment (not illustrated),diplexer 510 may include a third sub-diplexer at reader junction 507,the third sub-diplexer being characterized by two passbands, one of thetwo passbands allowing signals whose channel frequencies fall withinchannels 1-4 to pass along waveguide/transmission line 520 tosub-diplexer 510 a and the other of the two passbands allowing signalswhose channel frequencies fall within channels 5-8 to pass alongwaveguide/transmission line 520 to sub-diplexer 510 b.

As mentioned above, SAW circuits can provide narrowband filteringfunctions as the basis for a centralized or distributed diplexer system.Another approach is to use narrow-band antenna elements such that thefiltering function is provided by the antenna. For example, microstrippatch antennas can closely resemble parallel RLC resonant circuits.Other types of antennas closely resemble series RLC resonant circuits.Although the antenna filter response is not usually as narrow as may beobtained using SAW devices, there are some advantages, such as fewerparts, smaller size, and lower mass, that may be realized when tighterchannelization is not required. Several such embodiments usingnarrow-band antenna elements are described below. A narrow band antennais thus characterized by a passband corresponding to a range offrequencies that the antenna will pass or transmit (the narrow bandantenna is designed to filter out frequencies falling outside of therange of frequencies).

The FM RFID systems described above may generally be considered to bedesigned as sparse zone RFID technology, although they are notnecessarily so limited. Below, FM RFID systems generally designed fordense zones (e.g., smart shelves, smart enclosures, and smart surfaces)are described, although again they are not necessarily so limited.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 6. FIG. 6 is a schematic diagram of an FM RFID system600 for a smart shelf 601, the system including a reader 605, a diplexer610, and a plurality of open (or at least partially open) transmissionlines 615, wherein the diplexer 610 is implemented in a distributedfashion similar to that shown in FIG. 2. System 600 (FIG. 6) may thus bethought of as a modification of system 200 (FIG. 2), in particular, asapplied to a smart shelf. A main difference between system 600 andsystem 200, in terms of structural components, is that system 600employs open transmission lines 615 instead of the antennas 215 ofsystem 200. Accordingly, in system 600 an electromagnetic signal istransmitted along a transmission line 615 in response to and at thechannel frequency of the RFID signal distributed thereto rather than thesituation in system 200 wherein an electromagnetic wave is transmittedby an antenna 215 in response to and at the channel frequency of theRFID signal distributed thereto. The electromagnetic signal transmittedalong transmission line 615 effectively reaches RFID tags by near-fieldcoupling or radiation from transmission line 615. The term “(at leastpartially) open” is used here to mean that the transmission lines 615are not entirely electromagnetically shielded (the electric fields alongthe transmission lines 615 are not entirely blocked), so that at leastsome electromagnetic energy or radiation from the transmission lines 615may couple to nearby RFID tags, either by near-field coupling orradiation. In some embodiments, the transmission lines 615 may becovered by a non-RF-opaque material, i.e., a material that permits RFelectromagnetic radiation to pass through it.

As seen in FIG. 6, frequency multiplexing is used to transmit signals todifferent regions (zones 1-4) on smart shelf 601, via respectivetransmission lines 615. As discussed above, a (micro)processor couldestimate the location of an RFID tagged item on smart shelf 601according to the responding channel or the relative signal strengths ofthe responding channels. That is, the processor would determine that theitem is located in the coverage region (zone) of the one of thetransmission lines 615 that receives the response signal from the item'sRFID tag (most strongly). Thus, the smart shelf system 600 of FIG. 6could distinguish the location of the tagged item (ID3) as being withinone of the four coverage zones 1-4 (specifically, coverage zone 2, inthe example illustrated in FIG. 6) on smart shelf 601 when thedistributed network of transmission lines 615 is fed from a single port606 on the reader 605. Many readers have the ability to switch betweenfour ports. Thus, by using the distributed network shown in FIG. 6 inconjunction with four ports, reader 605 could distinguish up to 16different regions (zones) on smart shelf 601. Much finer resolutioncould be achieved using narrower filters to isolate individual channels.For example, in the United States, as regulated by the FederalCommunications Commission, the EPCglobal Class 1 Generation 2 protocolmay employ frequency hopping spread spectrum over at least 50 channels.

System 600 is shown with waveguide 620, quarter-wave impedancetransformers 625, connections 622, filters 630, interfaces 635, andinterface spacings (m₁L, m₂L, m₃L, n₁L, n₂L) as in system 200 of FIG. 2(the lines shown between respective quarter-wave impedance transformers625 and respective filters 630, analogous to elements 222 in FIG. 2,represent mere connections, not physical lengths of feed/transmissionline). The description of analogous, like-numbered elements in system200 of FIG. 2 is applicable to these elements of system 600 of FIG. 6.Alternate arrangements in this regard (e.g., such as discussed above)are also possible for system 600.

We turn now from smart shelves (nominally two-dimensional regions) tosmart enclosures (nominally three-dimensional regions).

For purposes of this disclosure, smart enclosures may be thought of asRF cavities. While the discussion below refers to “drawers,” it will beunderstood that the smart enclosures discussed here may be any kind ofthree-dimensional container or any three-dimensional region, regardlessof whether it is (1) physically (a) completely enclosed or (b) ratherpartly enclosed and partly open, and regardless of whether it is (2)electromagnetically (a) completely enclosed (i.e., shielded) or (b)rather partly enclosed (shielded) and partly open (not shielded).Furthermore, the FM RFID systems, apparatuses and methods describedherein for application to smart enclosures are also applicable tothree-dimensional regions of space that are not necessarily enclosures,i.e., that may be open/unbounded or at least substantially open. Withregard to smart enclosures, FM RFID may provide fine localizingresolution as described above as well as radiation diversity,polarization diversity, and reduced volume requirements. One of thedesign goals of a feed system for a smart enclosure is to have the feedcircuit volume not detract appreciably from the usable storage volume ofthe enclosure. This design goal, however, typically conflicts with thedesire to cover the full bandwidth allocated to an RFID protocol. Forexample, in order to cover the full EPCglobal Class 1 Generation 2bandwidth, microstrip patch antennas typically have to be made fairlythick, e.g., 1 to 3 cm. (microstrip patch antennas become wider inbandwidth as their thickness increases). Furthermore, in order toachieve a desired read accuracy near 100% when the drawer contains manytagged items or contains items with substantial levels of liquid orconductive materials, it is typical to place two to four antennas aroundthe perimeter of the drawer. In addition to overcoming obscuration oftags and signal attenuation, the diversity achieved with multipleantennas can avoid problems with natural nulls in the modal fielddistribution as well as problems with polarization. The desire to havemultiple feed antennas throughout the drawer, however, not onlysubtracts from the storage volume, but also places requirements on thereader to switch among the multiple antennas. Because many readers haveintegrated four-port switches, operation with a single drawer does notusually constitute a severe restriction. However, for large drawers, orsystems of drawers, switch limitations can become problematic.

The distributed feed systems discussed above in the context of sparsezone RFID and smart shelves can be applied to smart enclosures toimprove on various limitations of smart enclosures such as thosediscussed above.

In this regard, an alternative embodiment, or set of embodiments, is nowdescribed with reference to FIG. 7. FIG. 7 is a schematic diagram of anFM RFID system 700 for a smart enclosure 701, the system including areader 705 and a plurality of narrow band reader antennas 715. Smartenclosure 701 is a three-dimensional enclosure, having a height a, alength b, and a depth c, as shown by the illustrated x, y and z axes.For brevity, the implementation details in the distribution line (fromreader 705 to antennas 715) are not shown in FIG. 7, given that one ofordinary skill in the art will understand this aspect of system 700 inview of the remainder of this disclosure. Narrow band antennas 715 maybe microstrip antennas and, more particularly, patch antennas. Whileeight antennas 715 are shown, a smaller or larger number of antennas 715may be employed. In some embodiments, one or more of antennas 715 may berotated relative to other(s) of antennas 715. For example, one or moreantennas 715 of the group of antennas 715 shown on the left wall ofenclosure 701 may be rotated in the plane of the left wall of enclosure701, which is the x-z plane (i.e., the plane formed by the x and zaxes), and one or more antennas 715 of the group of antennas 715 shownon the right wall of enclosure 701 may be rotated in the plane of theright wall of enclosure 701, which is parallel to the x-z plane. Suchrotation may be, e.g., by 90 degrees or another angle. Also, althoughantennas 715 are illustrated in a certain form, antennas 715 may employalternate narrow band designs, or a combination of two or more differentnarrow band designs, in order to optimize the amplitude and polarizationcoverage throughout the enclosure volume. In system 700, a single readerport 706 may feed all eight antennas 715, thus allowing other ports (notshown) to be connected to other drawers (not shown) or to antennas (notshown) on other surfaces of the same enclosure. Nonetheless, multiplereader ports 706 may be employed in system 700.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 8. FIG. 8 is a schematic diagram of an FM RFID narrowband antenna and feed system 800. System 800 may be used, e.g., forsmart enclosure 701, or for another type of region such as athree-dimensional region of space that is at least substantially open.System 800 represents one possible way of many in which the details ofsystem 700 may be elaborated. System 800 includes a reader (not shown inFIG. 8 but shown generally as reader 705 in FIG. 7) and at least onenarrow band reader antenna 815. As shown in FIG. 8, system 800 alsoincludes trunk line 820 and feed lines 821, 822 and 823. Feed lines 822and 823 feed a signal from either end of trunk line 820 to a respectiveone of antenna 815. Trunk line 820 feeds a signal from feed line 821 tofeed lines 822 and 823. Feed line 821 feeds a signal from a reader port(not shown in FIG. 8 but shown generally as reader port 706 in FIG. 7)to trunk line 820. Narrow band antennas 815 may be microstrip antennasand, more particularly, patch antennas. The number of antennas (andassociated feed lines) may vary from that illustrated in FIG. 8, asseen, for example, in FIG. 7 (showing four antennas operativelyconnected to an associated feed line).

Using the illustrated exemplary case of two narrow band antennas 815, afirst narrow band antenna 815 (e.g., shown at left) may be characterizedby a first passband, the first passband corresponding to a first rangeof frequencies, and the second narrow band antenna (shown at right) maybe characterized by a second passband, the second passband correspondingto a second range of frequencies, where the second range of frequenciesdiffers from the first range of frequencies, such that the secondpassband differs from the first passband. The second passband may differfrom the first passband, i.e., the second range of frequencies maydiffer from the first range of frequencies, in either of the followingtwo ways: (1) the first and second passbands may be overlapping (i.e.,the first range of frequencies and the second range of frequencies mayinclude one or more but not all frequencies in common) or (2) the firstand second passbands may be non-overlapping (i.e., the first range offrequencies and the second range of frequencies may be mutuallyexclusive, not having any frequencies in common). The first narrow bandantenna 815 is configured to transmit a first electromagnetic wave inresponse to and at the channel frequency of the RFID signal fed theretoby the associated first feed line (shown at left), if the channelfrequency of the RFID signal falls within the first passband, and thesecond narrow band antenna 815 is configured to transmit a secondelectromagnetic wave in response to and at the channel frequency of theRFID signal fed thereto by the associated second feed line (shown atright), if the channel frequency of the RFID signal falls within thesecond passband. (In the case that the first and second passbandsoverlap, each of the narrow band antennas 815 would transmit anelectromagnetic wave in response to and at the channel frequency of theRFID signal fed thereto if the channel frequency of the RFID signalfalls within the overlapping region of the first and second passbands.)As mentioned above with reference to other embodiments and as holdsgenerally throughout this disclosure, the channel frequency of the RFIDsignal generated by the reader 805 and fed by the feed lines to therespective antennas 815 may vary over time, e.g., according to an FHSSscheme. The term “first feed line” may be used to refer to the entirefeed line from reader port to first (e.g., shown at left) narrow bandantenna 815, including feed line 821, left portion of trunk line 820,and feed line 822; the term “second feed line” may be used to refer tothe entire feed line from reader port to second (e.g., shown at right)narrow band antenna 815, including feed line 821, right portion of trunkline 820, and feed line 823. Each of the terms “first feed line” and“second feed line” may also be used to refer to a portion of thoserespective entire feed lines running from the reader port to therespective antenna. Similarly, the term “feed line” may be used to referto an entire feed line or to a portion of the respective feed linerunning from the reader port to the corresponding antenna.

Antennas 815 may be of sufficiently narrow band that their correspondingbandwidths do not overlap significantly, with the antenna centerfrequencies being close enough that half-wavelength (λ/2) lines (betweenantenna branch and reader port, along trunk line 820) andquarter-wavelength (λ/4) lines (feed lines 822, 823) are similar inlength at the two center frequencies of the antennas. Both patchantennas 815 may be edge-fed with microstrip lines of characteristicimpedance Z₀₁, and the patch input impedance at the edge of each patchantenna 815 may be Z_(e), as shown. Those skilled in the art, now havingthe benefit of this disclosure, would recognize that the input impedanceat the ends of the λ/4 lines (feed lines 822, 823) opposing the patchantenna 815 ends is given by Z₀₁ ²/Z_(e), where λ is the effectivewavelength of the microstrip line, and the effective wavelength is thephysical distance that corresponds to one wave cycle on the microstripline. Hence, when the frequency is sufficiently out of band for eitherpatch antenna 815, the edge impedance will appear close to 0, and theimpedance at the other ends of the quarter wavelength lines (feed lines822, 823) will appear to be infinite, or large enough to present aneffective open circuit at the ends of trunk line 820 where theyinterface with the respective quarter-wave lines (822, 823) (the furtherthe frequency is out of band, the more closely the edge impedance willapproximate 0 and the more closely the impedance at the other ends ofthe quarter wavelength lines 822, 823 will approach infinite impedance).The half-wavelength line (trunk line 820) will result in this apparentlyinfinite impedance being presented at the junction with feed line 821;hence, the out-of-band patch antenna 815 (e.g., shown at left) willappear as an open circuit within the operating band of the other patchantenna 815 (shown at right), and vice versa. For the in-band side, theantenna edge impedance must be transformed such that the input impedanceZ_(in) to the circuit in FIG. 8 is matched to the impedance of thetransmission line (not shown) that connects to the reader (not shown inFIG. 8 but shown generally as reader 705 in FIG. 7). The impedance atthe intersection of feed line 820 and 822, 823, in the direction of thein-band antenna, is also Z₀₁ ²/Z_(e), and at the in-band side, Z_(e) issubstantially greater than zero. The impedance at the intersection offeed line 821 and trunk line 820 is likewise (very nearly) Z₀₁ ²/Z_(e)in the direction of the in-band antenna. The length and characteristicimpedance Z₁ of transmission line 821 can be selected to furthertransform the impedance to the desired impedance Z_(in). Alternatively,Z₀₁ can be selected such that the transformed impedance Z₀₁²/Z_(e)=Z₁=Z_(in), such that no further transformation is required tomatch the in-band side to the reader impedance. In this manner, apredominance of power entering the input port in FIG. 8 is directedsubstantially to either the left or right antenna 815 according to thechannel or frequency of operation at that time. Assuming Z₀₁ and Z₀₂ areselected appropriately, the impedance of the in-band patch antenna canbe readily matched to the input impedance Z_(in). Z₀₂ may be selected tominimize line loss. Those skilled in the art, now having the benefit ofthis disclosure, would recognize that there exist multiple techniques toimpact the impedance at the antenna feed point. For example, instead offeeding the line at the very edge of the patch, the feed point can beinset into the patch to reduce the resonant impedance, Z_(e). Thus,multiple design degrees of freedom are available to match the impedanceand also to establish the resonant frequencies and bandwidths of the twoor more patch antennas 815.

It should be noted that the details shown in FIG. 8 exemplify one groupof embodiments of many possible. For example, as discussed previously(FIG. 2), the quarter-wavelength lines could be odd-integer multiples ofa quarter wavelength, and the half wavelength lines could be integermultiples of a half wavelength. For other antennas that can be modeledby a series resonant circuit, the lines 822, 823 would be integermultiples of a half wavelength instead of a quarter wavelength. Also,although transmission lines are illustrated as linear, they mayincorporate bends or meanders in order to satisfy geometric constraints,such as desired antenna spacing.

While the embodiments described thus far may provide spatial diversityby, for example, pointing the antenna in FIG. 8 in different directions,embodiments described further below may also provide polarizationdiversity. It will be noted that the arrangements described hereinprovide polarization diversity to the reader antenna.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 9. System 900 shown in FIG. 9 may be understood as avariant of system 800 shown in FIG. 8, with system 900 providingpolarization diversity unlike system 800. FIG. 9 is a schematic diagramof an FM RFID narrow band antenna and feed system 900, including areader (not shown in FIG. 9 but shown generally as reader 705 in FIG. 7)and a plurality of narrow band reader antennas 915. System 900 may beused, e.g., for smart enclosure 701, or for another type of region suchas a three-dimensional region of space that is at least substantiallyopen. As with system 800, narrow band antennas 915 may be microstripantennas and, more particularly, patch antennas. The number of antennas(and associated feed lines) may vary from that illustrated in FIG. 9, asseen, for example, in FIG. 7. Similar to antennas 815, in system 900 thetwo antennas 915 may be designed such that their operating bandwidthsare adjacent, but do not completely overlap. As with system 800 of FIG.8, system 900 also includes trunk line 920 and feed lines 921, 922 and923, interconnecting the analogous elements as in system 800, and withsimilar half-wavelength spacing between antenna branch and reader port,as shown. However, feed line 923 of system 900 differs from feed line823 of system 800 in that feed line 923 feeds a (first) edge of rightside patch antenna 915 orthogonal to a (second) edge of left side patchantenna 915 fed by feed line 922, whereby the radiating edges of rightside antenna 915 (the vertical edges) are orthogonal to the radiatingedges of left side antenna 915 (the horizontal edges), and hence, theradiated fields from the respective antennas 915 will possess orthogonalpolarizations, one vertical and the other horizontal. As with system800, the antennas 915 are characterized by different, possiblyoverlapping passbands. In regions where the passbands overlap, the setof antennas 915 may radiate such that the fields combine, and thepolarization may become elliptical, including the special case in whichthe polarization is linear diagonal, i.e., the polarization vector is at45 degrees to the polarization of either of the antennas 915. Similar towaveguide 823, waveguide 923 is a quarter-wavelength or an odd integermultiple of a quarter-wavelength, even though it might be characterizedby one or more right angles, curves, or mitered bends in order to feedthe orthogonal edge. For other types of antennas that are well modeledby a series RLC circuit, the waveguides 922, 923 are even integermultiples of a quarter wavelength. Although the two antennas 815 looksimilar, and the two antennas 915 look similar, as is known to thoseskilled in the art, now having the benefit of this disclosure, thelength of the antenna is a factor in establishing the resonantfrequency, and often only small differences in length will offset theresonant frequency, particularly for narrowband patch antennas.

An alternative embodiment, or set of embodiments, is now described withreference to FIG. 10. FIG. 10 is a schematic diagram of an FM RFIDnarrow band antenna and feed system 1000, including a reader (not shownin FIG. 10 but shown generally as reader 705 in FIG. 7), a microstrippatch reader antenna 1015 fed by two feed lines 1022, 1023. Trunk line1020 connects feed lines 1022, 1023 to a reader (again, not shown inFIG. 10 but shown generally as reader 705 in FIG. 7). System 1000 may beused, e.g., for smart enclosure 701, or for another type of region suchas a three-dimensional region of space that is at least substantiallyopen. Like system 900, system 1000 provides polarization diversity, butunlike system 900 system 1000 uses one microstrip patch antenna 1015 toachieve polarization diversity of the generated RFID signal of thereader. In each of systems 800 and 900, small differences between thelengths L₁ and L₂ of the two respective patch antennas 815, 815 (or 915,915) are typically used to offset the resonant frequencies of the patchantennas (i.e., patches) relative to each other, although other methods,such as perturbations to one or both patches, can also be used to offsetthe resonances (the length here mentioned is defined as the distancefrom the fed edge to the opposing edge of the patch). The orthogonalpatch dimension, defined as the width (W₁ or W₂) affects the bandwidthof the patch but has little effect on the resonance. So, the widths (W₁and W₂) of patch antennas 815, 815, in system 800 and the widths ofpatch antennas 915, 915 in system 900 can be set to establish thedesired patch bandwidth. However, in system 1000, the width W and thelength L of patch antenna 1015 are set to have magnitudes different fromeach other so as to create two orthogonal resonances in patch antenna1015 that are sufficiently separated that their respective operatingbandwidths do not completely overlap. For example, with a laminatecharacterized by a relative permittivity of approximately 3 and alaminate thickness of 0.175 inches, one dimension of the patch antennamight be 3.630 inches, and the orthogonal dimension might be 3.505inches. Feed lines 1022 and 1023 may each have a length equal to an oddinteger multiple of a quarter wavelength. The length of feed line 1022may but need not be equal to the length of feed line 1023. For example,the length of feed line 1022 may be an odd integer multiple ‘m’ of aquarter wavelength, and the length of feed line 1023 may be an oddinteger multiple ‘n’ of a quarter wavelength, where m and n may but neednot be the same odd integer. The “length” of feed line 1022 or 1023refers to the distance from the junction with trunk line 1020 to thejunction with antenna 1015. With these length constraints, the feedlines 1022, 1023 serve as impedance transformers, and hence preventsignals having out-of-band frequencies from traveling down the wrong oneof feed lines 1022, 1023, similarly as discussed above with respect topreviously described embodiments. In the case of system 1000, antenna1015 generates electromagnetic waves having different polarizations, inresponse to receipt, respectively, of the two signals fed fromrespective feed lines 1022, 1023. The different polarizations may becharacterized by different orientations, e.g., they may be orthogonal toeach other. Further, the different polarizations may be vertical andhorizontal, respectively, or vice versa. Over spectrum regions in whichthe passbands of the two modes overlap, the antenna may exhibit yetdifferent polarizations. In some embodiments, feed line 1022 and feedline 1023 are of the same electrical length such that the phase of theRFID signal associated with propagation thereof along each of feed line1022 and feed line 1023 is the same, and the first passband and secondpassband overlap such that the electromagnetic wave generated by antenna1015 in response to a signal from feed line 1022 having a firstpolarization characterized by a first (e.g., linear) orientation and theelectromagnetic wave generated by antenna 1015 in response to a signalfrom feed line 1023 having a second, different polarizationcharacterized by a second, different (e.g., linear) orientation jointlyresult in a polarization characterized by a (e.g., diagonal) orientationbetween the first orientation and the second orientation. Suchintermediate (e.g., diagonal) orientation provides additionalpolarization diversity.

An alternative embodiment, or set of embodiments, is now described withreference to FIGS. 11A and 11B. FIGS. 11A and 11B are schematic diagramsof FM RFID narrow band antenna and feed systems 1100A and 1100B,respectively, each system including a reader (not shown in FIGS. 11A and11B but shown generally as reader 705 in FIG. 7) and a microstrip patchreader antenna 1115A or 1115B, respectively, fed by a feed line 1122A or1122B, respectively. Systems 1100A and 1100B may be used, e.g., forsmart enclosure 701, or for another type of region such as athree-dimensional region of space that is at least substantially open.Like systems 900 and 1000, systems 1100A and 1100B provide polarizationdiversity, but unlike those systems, each of systems 1100A and 1100Buses one microstrip patch antenna 1115A or 1115B, respectively, pairedwith one feed line 1122A or 1122B, respectively, to achieve polarizationdiversity of the generated RFID signal of the reader, i.e., a “singlefeed microstrip patch” design. Unlike feed lines 921, 922 and 923 insystem 900 and feed lines 1022 and 1023 in system 1000, because power isdivided between two modes internal to the patch 1115A or 1115B insystems 1100A and 1100B, feed lines 1122A and 1122B are not constrainedto an odd integer of a quarter wavelength for the purpose of creating ahigh impedance that reduces or blocks power from coupling to the antenna1115A or 1115B, respectively. However, feed lines 1122A and 1122B may bea quarter wavelength line used to match impedances as needed.

Each of antennas 1115A and 1115B is a single feed microstrip patchdesign that can excite two modes, one that radiates a substantiallylinear first polarization, and another that radiates a substantiallylinear second polarization, the two polarizations being characterized bydifferent orientations. The two polarizations may be orthogonal to oneanother, e.g., horizontal and vertical. This patch design is frequentlyused to create two modes in substantially equal proportions with respectto magnitude with one mode excited approximately 90 degrees ahead of, orbehind, the other (e.g., orthogonal) mode. The formation of twoorthogonal polarizations that are separated by approximately 90 degreesin phase creates waves that are circularly polarized, or at leastelliptically polarized waves that are very nearly circular inpolarization. The formation of two modes that are nearly equal inamplitude but separated by 90 degrees in phase is achieved by perturbinga substantially square conductive patch such that the mode resonancesare slightly displaced. As those of ordinary skill in the art wouldunderstand, now having benefit of this disclosure, there are multipleways to achieve this effect, i.e., a multitude of single-feed approachesto achieve a circularly polarized variant of the single feed multistrippatch antenna. While a single-feed circularly polarized patch providesvalue in polarization diversity for many applications, for at least someRFID applications there may also be disadvantages. Embodiments describedherein include additional aspects or features in order to optimize RFIDlinks when the channel frequency varies over time. More specifically,these other embodiments realize a multi-linear polarization antennarather than a circularly polarized antenna. The term “multi-linearpolarization” is used to mean highly elliptical polarization in whichthe tilt angle of the polarization vector varies as a function offrequency, as opposed to simply dual polarization, as described furtherbelow. The tilt angle, or polarization tilt angle, is the angle that thepolarization vector makes with an axis of a coordinate system. The term“highly elliptical” is used to mean an ellipse with an axial ratio of 3dB or greater in some embodiments, and in other embodiments, it means anellipse with an axial ratio greater than or equal to 5 dB.

Techniques, in a single feed microstrip patch design, to perturb themodes such that one resonates at a slightly different frequency than theother can be classified into at least two different types: those thatalter an aspect along a diagonal dimension of the patch antenna and havea feed line aligned with a horizontal or vertical dimension, oftenreferred to as “Type A”, and those that alter an aspect along ahorizontal or vertical dimension (i.e., corresponding to the dimensionsof W or L) with a feed line along a diagonal line, often referred to as“Type B”. Another way of describing these two types is that in Type Athe configuration of the patch is altered in a diagonal direction andthe patch has a feed aligned with a width or length direction, and inType B the configuration of the patch is altered in a direction of thewidth or length and the patch has a feed aligned with a diagonaldirection. In Type A, the alteration of the configuration of the patchmay be, e.g., in a central region or at a corner of the patch. In TypeB, the alteration of the configuration of the patch may be, e.g., in acentral region of the patch, in a central region of a side of the patch,or along an entire side of the patch. In either Type A or Type B, thealteration of configuration may be a truncation (also referred to as“chamfering”) or cutout, or, oppositely, an extension or tab (extendingoutward of the patch). An example of the first approach (Type A),illustrated in FIG. 11A, is to truncate the corners of a square patch1115A, as shown by reference numerals 1119A, with the degree oftruncation being sufficient to separate the mode resonances beyond thatused for circular polarization. Thus, W and L would be equal in thisapproach. An example of the second approach (Type B), corresponding to aperturbation along a dimension W or L, illustrated in FIG. 11B, is touse a rectangular patch 1115B whose shape is relatively close to square.That is, the length L of the patch would differ slightly in magnitudefrom the width W of the patch 1115B. Also, in this approach, the feedline 1122B intersects the patch 1115B at a corner region 1119B of thepatch 1115B. In contrast to single feed circularly polarized antennasthat may use a similar architecture, in system 1100B the differencebetween length L and width W is, for a set patch thickness, greater inorder to impart greater separation between the modal resonantfrequencies, thus eliminating the condition for circular polarization.Thus, in system 1100B, the width W and the length L of patch antenna1115B may be set to have magnitudes different from each other, as withpatch antenna 1015 of system 1000.

The unloaded quality factor (Qo) of a single mode determines, in part,the amount of patch perturbation to produce the multi-linearpolarization response. The quality factor Qo also determines theperturbation to achieve the circular polarization response for thesingle feed circularly polarized patch, and for both designs (i.e.,circularly polarization and multi-linear polarization) the qualityfactor is assumed to be the same for both modes. A primary difference isseen, however, in the relationship between Qo and the magnitude of theperturbation. A perturbation (or perturbation area, also referred to asa delta area), Ds, is defined as an area either added to or subtractedfrom the top (i.e., non-ground plane) side metallization of a patchantenna. An area S is defined as the area of the unperturbed top sidemetal pattern of a patch. For the conventional Type A circularlypolarized patch, the ratio of Ds to S equals 1 divided by the product of2 and Qo (that is, Ds/S=1/(2*Qo). For Type B designs, the conventionalcircularly polarized patch prescribes the ratio of Ds to S as 1 dividedby Qo (that is, Ds/S=1/Qo) (see, e.g., Microstrip Antenna DesignHandbook, Prakash Bhartia, et al, 2001, pp. 503-515 Artech House). Forthe circularly polarized patch designs of both Type A and Type B, Qo istypically selected based on the desired axial ratio bandwidth, theimpedance bandwidth, and the desired efficiency, although the axialratio bandwidth is usually the most important factor. In other words, ifthe selected Qo satisfies the axial ratio bandwidth, it usually alsosatisfies the impedance bandwidth and the efficiency requirements. Inboth Type A and Type B designs, for the circularly polarized patchdesign resulting from the selected Qo and the resulting perturbationarea, Ds, the normalized magnitudes of the impedances resulting fromeach mode are approximately 0.707 of the respective peak normalizedmagnitude values at the mid-band frequency at which they cross. In otherwords, if each mode has a normalized peak impedance response of 1.0 atthe resonant frequency of that mode, the two modes cross at a relativeimpedance magnitude of 0.707 near the center frequency. The peakresistive, and impedance, values occur at the center resonancefrequencies for each mode, respectively. In contrast, an exemplaryembodiment for a single feed multi-linear polarization design isdescribed next.

To convey the design process of the multi-linear polarized variant ofsingle feed microstrip patch design, the following parameters aredefined:

Bo: fractional operational bandwidth of the system

Bv: fractional 2:1 VSWR bandwidth of each individual mode, assumed to bethe same for each of the two individual modes

fa: center frequency of lower resonant mode

fb: center frequency of upper resonant mode

fc: center frequency of operating bandwidth; also midpoint between moderesonances

sigma: ratio of mode separation to the product of the fractionaloperational bandwidth and tire center frequency, fc

X: the ratio of the fractional VSWR mode bandwidth to the operationalbandwidth

Ds: area of perturbation of top side patch metallization

S: area of top side patch metallization

Using these parameters, for a Type B single feed patch, the perturbationfor the multi-linear design can be expressed in terms of the fractionaloperational bandwidth as:Ds/S=(2sigma Bo)/(2+sigma Bo)  Eqn 1

Equivalently, the perturbation ratio Ds/S can be expressed in terms ofthe fractional 2:1 VSWR bandwidth as:Ds/S=(2sigma Bv)/(2X+sigma Bv)  Eqn 2

The latter equation can be equivalently expressed in terms of Qo:Ds/S=(2sigma)/(2sqrt(2)X Qo+sigma)  Eqn 3

It should be noted that X and Bv in Eqn 2 are related quantities, andlikewise X and Qo in equation 3 are related quantities, such thatsetting one quantity also establishes the other. However, the equationsare useful for comparing the increased magnitude of perturbation ratioDs/S for a single feed multi-linear polarized patch compared to that forcircularly polarized patches, which is often expressed in terms of Qo.

For a Type A single feed multi-linear polarized patch antenna, theequations establishing the perturbation ratio Ds/S are expressed asfollows:Ds/S=(sigma Bo)/(2+sigma Bo)  Eqn 4Ds/S=(sigma Bv)/(2X+sigma Bv)  Eqn 5

The latter equation can be equivalently expressed in terms of Qo:Ds/S=(sigma)/(2sqrt(2)X Qo+sigma)  Eqn 6

As explained previously, the multi-linear design may separate the modesto a greater extent compared to the circularly polarized design. Also,because the multi-linear design generally utilizes higher Qo modes thanthe circularly polarized designs, the peak resistance corresponding tothe former design may be greater than the latter design. For a goodimpedance match, it is typically desired that the variation inresistance not be too great within the operational band. Further, inorder to achieve a high polarization aspect ratio (i.e., the ratio ofthe major to minor axis of the polarization ellipse) at the midbandpoint, it is desired that the voltage phase difference between the twomodes be substantially different than ±90 degrees at the midband point.For perfect linear polarization, the phase difference would be 0 or 180degrees, although good near-linear polarization performance can beachieved without reaching these extremes. As the modes are moved furtherapart, the phase difference separation from ±90 degrees is increased.So, greater separation of the modes toward the edge of the operationalbandwidth may improve both the impedance match throughout the band andthe linearity of the polarization at the midband point. If the mode bandcenters fb and fa are removal too far outside the operational bandlimits, the desired polarization response is not achieved. Inparticular, the polarization tilt axis, tau, does not varysubstantially, as will be discussed further below. Thus, for themulti-linear design, in order to achieve a good impedance match, a goodelliptical aspect ratio, and a wide range of polarization tilt angles,the mode center frequencies, fa and fb, are located toward the ends ofthe operational band, either just inside or just outside the operationalbands limits. The parameter sigma defines this separation as the ratioof the mode frequency separation (fb−fa) to the fractional operationalbandwidth:sigma=(fb−fa)/(Bo fc)  Eqn. 7

A value of sigma=1 places the lower and upper mode frequencies, fa andfb, respectively, to be coincident with the lower and upper limits ofthe operational bandwidth. In practice, values of sigma might beslightly larger than 1 or slightly less than 1 to balance requirementsfor polarization tilt, axial ratio, and impedance match. For example,sigma might be 1.3 or 0.8.

In an exemplary embodiment, a design is initiated by selecting Qoaccording to a desired fractional operating bandwidth, Bo. A value of Xis selected in order to meet impedance and polarization requirements. Ingeneral, a value of X approximately equal to 0.47 is found to permit agood impedance match as well as good polarization characteristics. Inother cases, discussed below, a different value of X might be used, forexample to meet an efficiency requirement. For this exemplaryembodiment, it is assumed that the value of Qo expressed by thefollowing equation:Qo=1/(sqrt(2)X Bo)  Eqn. 8

results in an acceptable value of efficiency when X=0.47. In this case,a value of sigma between 0.95 and 1.3 produces very good polarizationtilt angle and axial ratio results. For example, the value of sigma=1.3provides a polarization tilt angle that covers all but 5.4 degrees of afirst quadrant, and a minimum axial ratio of 5.8. The value ofsigma=0.95 provides a polarization tilt angle that covers a fullquadrant and about 15 degrees of a second quadrant, and a minimum axialratio of 3.2 dB over a quadrant. The final selection of sigma may bedictated according to satisfying requirements for the input impedancematch.

It should be emphasized that equation 8 is expressed in terms of afractional operating bandwidth, such that the results are extensible toa very wide range of operating bandwidths. In some extreme cases, theoperating bandwidth of the system may be so narrow that the Qo accordingto Eqn. 8 results in an unacceptably low efficiency. In such cases, alarger value of X may be used to result in a larger fractional VSWRbandwidth and hence lower Qo.

Also of significance, with single feed circularly polarized patchantennas, designers typically minimize Qo to broaden the axial ratiobandwidth of the patch. In fact, in many single feed circularlypolarized designs, the axial ratio bandwidth requirement drives thedesigner to use a low Qo much more than does the impedance bandwidth.Lowering the Qo is typically done by increasing the thickness of thepatch or decreasing the relative permittivity of the substrate betweenthe top and bottom metal layers, which necessitates an increase of theoverall metallized area. In contrast, with the multi-linear patchdesign, higher mode Qo's are typically desired as this results ingreater axial ratios (higher eccentricity, or degree of polarizationlinearity). For example, as described above, the fractional VSWRbandwidth Bv may be only, approximately, one half (0.47 in the precedingexample) of the operational bandwidth, and hence the Qo may beapproximately twice that of a Qo designed for the full operationalbandwidth. The higher Qo is typically achieved by reducing the patchthickness, which is usually also a trait for size and weightconsiderations. The higher Qo may also be achieved by increasing therelative permittivity of the substrate between the top and bottommetallization areas of the patch, which necessitates a reducedmetallization area for fixed resonant frequencies.

Increasing the patch thickness typically increases the patch efficiency,ignoring effects of surface waves. However, the thickness to achieveacceptable circular polarization over an operational bandwidth is oftengreatly in excess of the thickness for acceptable patch efficiency. Incontrast, the design of the multilinear polarized single feed patchantenna allows for the thickness to be reduced down to the desiredefficiency of the patch. In fact, due to the reduction of polarizationlosses afforded by the multi-linear polarization, in comparison to the 3dB one-way loss between a circularly polarized antenna and a linearlypolarized antenna, the designer may be afforded the design freedom toreduce the efficiency of the patch, and hence also further reduce thethickness of the patch.

The following example is specific to designing a Type B multi-linearpolarized single feed patch: We defined a delta area, Ds, as the productof (i) the difference between length and width and (ii) width, wherewidth is shorter than length. That is, Ds=(L−W)*W. We define an area, S,that is defined as W*W.

It is of interest to examine the difference in perturbation areasbetween the circularly polarized (CP) single feed designs and themulti-linear polarized single feed designs. For the Type B CP design,the perturbation is prescribed to be Ds/S=1/Qo. In contrast, for themulti-linear polarized single feed design, if it is assumed X isapproximately 0.5, that 1.4 Qo>>1 (which holds for all patches ofpractical interest), and that sigma is at least 1, then a lower bound onthe perturbation ratio may be expressed byDs/S>sqrt(2)/Qo  Eqn. 9

In this case, the perturbation area is at least 40% greater than thatassociated with the Type B CP design. In one or more embodiments,sigma=1.3, such that Ds/S is approximately 2 times the perturbationratio for the Type B CP design. Similar ratios may be found in comparingthe Type A versions of both designs. Because the mode separations aredetermined by tire perturbation ratios, it can be seen that the moderesonances in the multi-linear design are separated to a greater degreeby these same factors; that is, >40% or approximately 2 times theseparation in one or more embodiments compared to the CP designs. As anexample, for a laminate characterized by a relative permittivity ofapproximately 3 and a laminate thickness of 0.175 inches, one dimensionof a Type B patch might be 3.630 inches, and the orthogonal dimensionmight be 3.505 inches. The resulting unloaded quality factor (Qo) ofeither mode may be approximately 53 with a corresponding 10 dB returnloss bandwidth of 14 MHz, such that the resulting superposition of thetwo staggered modes can cover the entire band defined by the NorthAmerican (FHSS) implementation of the EPCglobai Class 1 Generation 2standard. As stated above, a delta area, Ds, is defined as the productof (i) the difference between length and width, and (ii) width, wherewidth is shorter then length. The delta area, divided by the area, S,approximately established by Equation 1, 2, or 3 above, results in aratio of 0.0366. In contrast, for single-fed circularly polarized patchdesigns, this ratio is prescribed to be 1/Qo, or in this case 0.019(see, e.g., Microstrip Antenna Design Handbook, Prakash Bhartia, et al,2001, pp. 503-515 Artech House). In other words, the type B designtaught herein calls, in this example, for a ratio of delta area dividedby area that is approximately 1.9/Qo, which results in an extension ofone dimension that is approximately 1.9 times greater than is taught forsingly-fed circularly polarized antennas with the same Qo.

For single feed patches of Type A, the condition for circularpolarization is that the delta area divided by the area is equal to 1/(2Qo), where in this case the delta area is the additional patch areaadded to or removed along the diagonal dimension. Following thetechnique for the Type B case with circular polarization would result ina delta area divided by area that is approximately 1/Qo.

To further underscore the differences in the design and functionality ofthe multi-linear polarized Type B single feed patch antenna compared tothe circularly polarized (CP) Type B single feed patch antenna, FIG. 20shows the normalized magnitude voltage responses of the two modes (upperand lower resonant modes) in a single feed circularly polarized patchantenna in which Qo=25. The x-axis is in units of MHz and corresponds toan operational RFID band from lower frequency 2004 at 902 MHz to anupper frequency 2005 at 928 MHz. The y-axis represents the normalizedmagnitude voltage response. The conditions for circular polarizationresult in a lower mode center frequency 2006 and an upper center modefrequency 2007. As required for the circular polarization condition ofthe single feed patch, the responses intersect at midband at point 2001with a value of approximately 0.707 relative to the peaks, which have anormalized value of 1.0. In practice the peaks of the two resonant modesmay not be exactly the same magnitude level. The phase difference (notshown) between the two modes at this point is approximately ±90 degrees,as it is for circular polarization. For perfect circular polarization,the amplitudes are equal, so the axial ratio of the antenna tends tobecome elliptical at the band edges, where the difference in amplitudesshown by arrows 2002 and 2003 is such that the voltage magnitude ratiois 1.9. However, because of the low Qo, the amplitude and phase areclose enough to the conditions for circular polarization (equalamplitude and ±90 degree phase difference) that the axial ratio does notexceed 6.3 dB even at band edges. If the patch Qo is increased to 53,and the condition for circular polarization is maintained by moving themodes closer together, the normalized magnitude voltage responses becomenarrower as shown in FIG. 21. The units of the x-axis in FIG. 21 arealso in MHz. Specifically, the mode center frequencies 2106 and 2107 areinside of the operational band limits 2104 and 2105. The crossing point2101 remains at a level of 0.707, as it should for the circularpolarization condition, and similarly the phase difference between thetwo modes (not shown) is approximately 90 degrees at this point. Theseparation 2102 at the lower operational band limit 2104 and theseparation 2103 at the upper operational band limit 2105 of theoperating band at the edges has increased to result in a magnitude ratioof approximately 2.4, which results in a higher axial ratio at the bandedges. This design would typically be considered under-designed in thesense that it does not achieve good axial ratio performance across theentire operating bandwidth. However, this sub-optimal performance issometimes used for single feed circularly polarized patch designs asincreased axial ratio bandwidth often comes at the expense of unsuitablylarge patches or increased cost. Although the under-designed CP singlefeed patch becomes linear at the band edges, it will be shown below thatthis under-designed CP single feed design is inferior to the specificmulti-linear design. In other words, the multi-linear design does notresult simply from applying the conditions for circular polarizationwith a Qo that is too high to obtain broad circular polarization acrossthe operating bandwidth. Rather, the multi-linear single feed designrequires a specific Qo relative to the operating bandwidth and properspacing of the two resonant modes.

FIG. 22 illustrates the normalized magnitude voltage responses of thetwo modes (upper and lower resonant modes) for the multi-linearpolarized single feed design when the single mode Qo=53 and sigma=1.3.The units of the x-axis in FIG. 22 are also in MHz. A lower moderesponse has a center frequency 2206 and an upper mode response has acenter frequency 2207. At midband, the magnitudes intersect at a point2201 with a voltage magnitude response of approximately 0.45.Furthermore, the phase (not shown) difference at midband is 125degrees—substantially different than the ±90 degrees for circularpolarization. So, even though the two magnitudes are equal at midband,the phase difference is such that the two modes together produce ahighly elliptical response (5.8 dB axial ratio), as opposed to acircular polarization response. At the lower operating band edge 2204and at the upper operating band edge 2205, the magnitude ratios shown byarrows 2202 and 2203, respectively, have increased to about 3.2, so thatgreater polarization linearity is achieved at the band edges of themulti-linear patch design compared to the circular polarization design.

One of the significant features of the multi-linear polarized singlefeed patch design is that the design results in a polarization vectorthat advances from horizontal to vertical as the frequency is increasedacross the operational band, and that at each increment, or step infrequency, the polarization response is characterized by an axial ratiothat is 3 or higher. In other words, the multi-linear polarized singlefeed patch produces polarization tilt angles, with respect to an axisreferenced to the patch, that spans at least nearly 90 degrees as thefrequency increases from the lower limit of the operational band to theupper limit of the operational band, and at each frequency step theaxial ratio is at least 3. The phrase “nearly 90 degrees”, as usedherein, means at least 70 degrees in some embodiments and greater than80 degrees in other embodiments. This multi-linear functionality is dueto the (i) gradual transition of radiation from one dominant mode at oneend of the operating band to the other mode at the other end of theoperating band, where “dominant” refers to the stronger voltage responsebetween the two modes; and to the (ii) relative phase response beingsubstantially different than ±90 degrees, preferably 0 degrees or 180degrees. The vector summation of the fields from the two modes resultsin a major axis of the polarization ellipse and a correspondingpolarization tilt angle that transitions across the span of polarizationtilt angles as the channel frequency transitions from one end to theother of the operational bandwidth. It is possible to design a singlefeed antenna in which the polarization response is highly elliptical,yet the polarization tilt angle does not cover the angular region of aquadrant as the frequency is shifted across the operational band. As anexample, the circularly polarized patch with Qo=53 from the previousexample is reconsidered. Typically, antenna designers select acircularly polarized antenna to control polarization loss incommunication links in which one end is served by a linearly polarizedantenna with unknown orientation. Single feed antennas are often chosenfor manufacturing ease and small size. However, broadband low axialratio performance, which requires a low Qo, is difficult to achieve withsingle feed patch designs, especially if the patch thickness is aconstraint. If a single feed design is applied in which the Qo is toohigh, a low axial ratio across the whole operational band is notpossible, as illustrated in the previous example displayed in FIG. 21 inwhich the Qo=53. In this case the polarization becomes linear at theband edges. However, the polarization vector does not sample the fullquadrant as a function of frequency, with a high axial ratio at eachfrequency within the operating band, as does the multi-linear design.This difference between the total angular extent of the circularpolarization design and that of the multi-linear polarization design isillustrated in FIG. 23, which is described next.

FIG. 23 shows two sets of polarization vector endpoints, 2303 (circularpolarization) and 2310 (multi-linear polarization), over a firstquadrant defined by positive x- and y-values in a Cartesian coordinatesystem with x-axis 2313 and y-axis 2314 (data set 2303 appears in thefigure as two separate data sets, but these are actually deemed part ofa single data set 2303; the explanation for this interpretation is givenbelow). The illustrated values along the x- and y-axes in FIG. 23represent normalized strength of the electric field (as normalizedvalues, they have no units); a given interval along the x-axisrepresents the same magnitude as the same interval along the y-axis. Inthe context of FIG. 23, “polarization vectors” are defined to besynonymous with the major axis polarization vector of an ellipticallypolarized electromagnetic field, and “polarization endpoints” aredefined to be the vector endpoints of that same major axis polarizationvector. In the figure, polarization vectors are denoted by referencenumerals 2305, 2306, 2315, 2317, and 2320 (note that each of referencenumerals 2305 and 2306 refers to two illustrated vectors). Each datasymbol in sets 2303 and 2310 represents an endpoint of a polarizationvector, although the polarization vectors having these endpoints are notshown, except for 2305, 2306, 2315, 2317, and 2320, so as not to detractfrom the legibility of the figure. For example, polarization vector 2315terminates at endpoint 2316. In each of sets 2303 and 2310, neighboringdata symbols represent a 1 MHz difference in frequency, and the totalspan of frequencies is from 902 MHz to 928 MHz, which is the operationalbandwidth in this example. Only endpoints corresponding to an axialratio greater than 5 dB are shown, as these are the points at whichpolarization loss is greatly reduced in transmission to a linearpolarized antenna. Data set 2303, denoted by circle symbols in the plot,corresponds to a single feed circularly polarized (CP) patch with Qo=53.The angle that a polarization vector makes with an axis, such as theangles 2319 and 2307 between the polarization vectors 2306,respectively, and the x-axis, is referred to as a polarization tiltangle. Similarly, polarization vector 2317 terminates at endpoint 2311with a polarization tilt angle 2318. Data set 2310, denoted by thesquare symbols, corresponds to the single-feed multi-linear (ML)polarized design. The mode spacing for the ML design is greater than theCP design as prescribed previously; that is, the mode spacing of the MLdesign is approximately twice as great as that of the CP design. Theradius, or magnitude of the polarization vectors in FIG. 23, isinsignificant; the difference in radius between the two data sets 2303and 2310 (or between the magnitude of CP polarization vectors 2305, 2306and the magnitude of multi-linear polarization vectors 2315, 2317, and2320) is introduced artificially and only serves to better distinguishthe two data sets 2303 and 2310. The span 2302, in which there are nodata points corresponding to the CP design, arises from the absence ofpolarization vectors over this angular span for the frequencies withinthe operating bandwidth. That is, over the span of frequencies thatdefine the operating bandwidth, the major axes of the polarizationellipses do not lie within this angular span for the CP design. Thereare two spans 2301 within the CP data set 2303 in which the axial ratiois greater than 5 dB, and the polarization vectors 2305 are shown at therespective angular ends of one of these spans 2301, and the polarizationvectors 2306 are shown at the respective angular ends of the other oneof these spans 2301. Thus, all the endpoints 2303 of the CP data setwithin the two spans 2301 represent polarization vectors withcorresponding axial ratios greater than 5 dB, although for the purposeof readability the polarization vectors having those endpoints 2303 arenot shown except for the four polarization vectors 2305 and 2306. Thetotal angular span covered by linear polarization, that is, the sum ofthe two spans 2301 (in this case, defined by axial ratio greater than 5dB), is approximately 22 degrees. In contrast, the total angular spancovered by the multi-linear polarization, indicated by the set of theendpoints 2310, covers almost the entirety of the quadrant, i.e., almost90 degrees.

It should be noted that the end points of the multi-linear span, such aspoints 2311 and 2312, correspond to the two end frequencies of theoperational band, 902 MHz and 928 MHz, respectively, in this example.So, whereas channel hopping at resolution steps finer than 1 MHz willresult in additional polarization vector endpoints within the interiorof the angular span, the outer limits 2311 and 2312 will not be exceededfor this particular design. In other words, in this specific designthere will be no tilt angle less than the angle 2318 formed by thevector 2317 to point 2311 (approximately 2.5 degrees) and the x-axis,and no tilt angle greater than the angle (not identified by referencenumber but shown to be approximately 87.5 degrees) formed by the vector2320 to endpoint 2312 and the x-axis. Thus, the polarization vectorscorresponding to endpoints 2311 and 2312 closely approach the horizontaland vertical axes, respectively. In contrast, larger gaps 2304 and 2319exist for the CP design data set. It should be noted that, for the CPdesign, these gaps 2304 and 2319 do not imply the absence of radiationwith a polarization vector aligned within regions 2304 and 2319. Rather,the polarization tilt angles within these regions will not also exhibitan axial ratio greater than 5 dB, and hence linear tags polarized in therange of the spans of the gaps 2304 and 2319 will entail higher minimumpolarization loss over the set of frequencies within the operating band,compared to the multi-linear design. While the difference in magnitudebetween gap 2304 (CP case) and gap 2321 (ML case), or between gap 2319(CP case) and gap 2318 (ML case) may not appear to be great, thisdifference is in fact highly significant. In other words, coverage(specifically, polarization tilt angle with axial ratio greater than 5dB) in one embodiment may be extended to as close to the coordinate axesas possible (or even beyond the axes). Similarly, the gap 2302associated with the CP case is a significant disadvantage, compared tothe ML case, because the ML case will exhibit reduced polarization lossover this angular region as well. Designing for polarization vectors(with polarization tilt angle with axial ratio greater than 5 dB) closeto the coordinate axes is a factor, not just for the coverage ofquadrant one, but more so for coverage into the neighboring quadrants,namely quadrants two and four. As is customary, quadrant two is definedas the set of coordinates [x<0, y>0]; quadrant three as coordinates[x<0, y<0], and quadrant four as coordinates [x>0, y<0]. The oscillationof the electric field creates the negative vectors to those shown inFIG. 23 such that quadrant three is covered to the same extent asquadrant one. For the multi-linear design, polarization coverage ofquadrants two and four is largely dependent upon the nearestpolarization vector in quadrants one or three. In this case, if thequadrant one polarization vectors are close to the x- and y-axes, as inthe preceding example with polarization vector 2317 and correspondingpolarization endpoint 2311, and the polarization vector 2320 andcorresponding polarization endpoint 2312, significant improvement in(i.e., reduction of) polarization loss is also achievable in quadrantstwo and four, although not to the same degree as quadrants one andthree. A worst case arises in which a corresponding antenna (i.e., theother end of the link) is polarized midway between the x- and y-axes inquadrants two or four (i.e., either 135 degrees or 315 degreescounterclockwise from the x-axis). In this case, assuming themulti-linear design achieves polarization vectors near the x- and y-axesin quadrants one and three, this worst case polarization loss is about 3dB. However, that loss occurs only at that one specific angle—everywhereelse the loss is less when the vector endpoints in quadrant one alignexactly with the x- and y-axes. With the small coverage gaps 2318 and2321 at the x- and y-axes, respectively, or in other words, with thesmall angles formed by polarization vector 2317 and the x-axis, and bypolarization vector 2320 and the y-axis, respectively, the worst casepolarization loss is slightly greater than 3 dB at angles 135 and 315degrees in quadrants two and four respectively, and up to a smallangular extent less than or greater than 135 and 315 degrees, the worstcase polarization loss tapers from slightly greater than 3 dB to 3 dB.For example, for the multi-linear design data illustrated in FIG. 23,the polarization vector 2320 ending at point 2312 is approximately 2.7degrees from the y-axis. Thus, the worst case polarization loss to a tagoriented at 135 degrees counterclockwise from the x-axis is 3.4 dB. At132.3 degrees and at 137.7 degrees the polarization loss would be 3 dB,and everywhere else within quadrant two the polarization loss isexpected to be less than 3 dB. In comparison, the theoretical 3 dB lossis precisely the loss expected everywhere in a link between a circularlypolarized antenna and a linearly polarized antenna. This example hasillustrated that the multi-linear single feed design does not arise byhappenstance in which only the Qo is increased. Rather, the multi-lineardesign, characterized by: (i) a highly elliptical polarization at eachchannel frequency; (ii) an ellipse major axis tilt angle that variesthroughout two quadrants (one and three or two and four) as a functionof channel frequency, and (iii) polarization vectors very near the x-and y-axes (i.e., a range of tilt angles that covers at least verynearly 90 degrees or possibly more than 90 degrees), must be carefullydesigned with both a measured selection of Qo and a corresponding modeseparation as prescribed above.

Although the example above was applied with respect to a specificcoordinate system, those skilled in the art, now having the benefit ofthis disclosure, will recognize that the coverage characteristics ofquadrants one and three may be interchanged, through design, withquadrants two and four. That is, in the preceding example, thepolarization coverage illustrated in FIG. 23 also extends to quadrantthree due to the oscillatory nature of the polarization vector ofelectromagnetic fields. Thus, the polarization loss characteristics inquadrant four are expected to be described similarly to those inquadrant two in the example above. So, quadrants one and three havestronger polarization coverage than quadrants two and four in thepreceding example. Those skilled in the art of antenna design, nowhaving the benefit of this disclosure, would recognize that themulti-linear polarized single feed patch antenna may be alternativelydesigned to provide the stronger polarization coverage in quadrant fouras opposed to quadrant one. Then, quadrants four and two would becharacterized by the stronger polarization coverage, and in thisalternative design the polarization coverage characteristics ofquadrants one and three would be interchanged with quadrants two andfour. In general, two of the quadrants will be characterized by strongerpolarization coverage than the remaining two.

The mode spacing, established by the variable “sigma” referenced above,and the ratio of the fractional VSWR mode bandwidth to the operationalbandwidth, represented by the variable “X”, can both be modifiedslightly to impact the axis gap angles 2318 and 2321. For example, avalue of sigma=1.2, instead of 1.3 as in the previous example (FIG. 22),results in axis gap angles 2318 and 2321 of only 0.05 degrees. Thistypically reduces the axial ratio at the center band. In this case, theaxial ratio at center band is reduced from 5.8 to 5.1 when sigma isreduced from 1.3 to 1.2. Similarly, Qo may be slightly reduced to lessenthe axis gap angles 2318 and 2321 at the expense of axial ratio at thecenter band. These slight variations would typically be traded inconsideration of impedance matching requirements in the design of themulti-linear single feed patch. For example, a more stringent impedancematching requirement might require sigma=1.3, whereas a less stringentimpedance matching requirement might permit sigma=1.2. In some cases, asufficiently lenient impedance matching requirement might permit sigmaequal to or slightly less than 1.0, in which case the axis gap angle2304 may vanish. In fact, polarization vector endpoint 2312 may crossover into quadrant two, further improving polarization coverage in thatquadrant.

There are a number of ways in which Type A and Type B patchperturbations may be implemented. A second type of Type A approach is tomodify system 1100B by employing cutouts of the conductive regions ofthe patch. Again, the cutout region is larger than the cutout regionused to achieve circular polarization, and in fact large enough torender the polarization highly elliptical or substantially linear.

With regard to single feed circularly polarized antenna systems, thefrequency bandwidth over which the patch is circularly polarized is verynarrow, in fact, significantly narrower than the overall impedance andradiation bandwidths. Outside of that narrow band in which the patch isnearly circularly polarized, the patch becomes linear horizontally andvertically polarized at opposite ends of the patch band, respectively.

There are at least two disadvantages to the conventional single feedcircularly polarized patch antenna, one of which applies to manyapplications, and the other of which is specific to many RFIDapplications. The first of these disadvantages is, as stated above, thatthe bandwidth over which the antenna exhibits good circular polarization(i.e., the axial ratio bandwidth) is quite limited, and is in fact foundto be significantly less than the impedance and radiation patternbandwidths of the antenna. To broaden the axial ratio bandwidth,designers may increase the thickness of the antenna such that the axialratio bandwidth fully covers the operating bandwidth. This increasedthickness sometimes results in unacceptably voluminous antennas.Designers may be able to reduce the relative permittivity of thesubstrate between the top and bottom metallization layers of the patch,but this type of change increases the cross-sectional area of the patch.Regardless of the difficulties with the single feed patch, thesimplicity of single feed designs has resulted in numerous deployments.

The second disadvantage of these single feed circularly polarized patchantennas is generally shared with the class of circularly polarizedantennas in many RFID applications, i.e., regardless of the design. MostRFID tags employ linear polarization, driven by the demand for low costand small-size tags and given the fact that circularly polarizedantennas are, in general, larger and more complex with higherfabrication cost. Because tag orientation is often unknown, and linearcross-polarization between a reader antenna and a tag antenna can resultin very high (theoretically infinite) link loss, and frequently to thepoint at which reader-tag communication is not possible, RFID readersfrequently employ antennas with circular polarization. However, whereasthe use of circularly polarized reader antennas at the reader preventsunlimited polarization loss, there is an associated nominal loss of 6 dBin the roundtrip link (3 dB polarization power loss from the readerantenna to the RFID tag, and another 3 dB polarization power loss fromthe RFID tag antenna back to the reader) because half of the power islost to polarization mismatch between circular and linear polarizations.

In view of the above disadvantages, this disclosure teaches a variationon the above approaches to a single feed circularly polarized patchantenna. This variation results in improved performance for RFID systemsin which the channel frequency is varied over time. According to thisvariation, the horizontal and vertical polarization mode centerfrequencies are separated to the low and high edges of the entireoperating band. For RFID systems employing FHSS, these edges wouldencapsulate the entire FHSS band. Thus, at one end of the FHSS band, theantenna operates as a horizontally polarized antenna, whereas at theother end of the FHSS band the antenna operates as a verticallypolarized antenna. Toward the middle of the band, the modes are roughlyequal in amplitude but without the ±90 degree separation that wouldconstitute a circularly polarized antenna. Instead, toward the middle ofthe band, the two modes tend to create a diagonal polarization (or anelliptical polarization with high aspect ratio) that lies between thehorizontal and vertical modes. In this manner, the 6 dB roundtrippolarization loss is avoided in RFID reader-to-tag communications. Inaddition, there is no need to increase the patch antenna thickness toincrease the axial ratio bandwidth. In fact, the requirement toband-limit each mode (frequency multiplex) drives the design to thinnerpatches.

The substantially multi-linear polarization described above is achievedby designing the patch antenna such that the resonant frequencies of thetwo modes are significantly further apart than is used for single-feedcircularly polarized (CP) antennas and by utilizing antenna unloadedquality factors that are significantly higher than typically employedfor single-feed circularly polarized antennas, as described above. Forexample, the aspect ratio of the nearly square design is greater thanfor the CP design, and the chamfered corner design as taught herein hasa larger chamfer compared to the CP design. Whereas in single-feed CPdesigns, the magnitude of the input impedance at the channel frequencyis typically 1/sqrt(2) below the peak impedance magnitude in order toachieve the ±90 degree phase separation, as taught herein the impedancemagnitudes of the two modes might cross at one-half or less of the peakimpedance magnitude.

The usual design choice in RFID communications is to employ a CP antennafor use with the reader in order to handle the arbitrary polarizationangle of the one or more tags, which typically have linearly polarizedantennas. The biggest disadvantage of this approach is that, nominally,3 dB of power is lost in the link from the reader to the tag, andanother 3 dB of power is lost in the link from the tag back to thereader. As an alternative to circularly polarized antennas, somepractitioners deploy systems in which the reader switches betweenmultiple linearly polarized antennas, typically between two and four.One disadvantage of this approach is that different antennas are usuallyemployed for spatial diversity; e.g., to look in different directions toensure optimal coverage so that tags are not missed. An antenna used forpolarization diversity cannot usually also provide adequate spatialdiversity. Moreover, an additional antenna must be used for each linearpolarization. If two antennas with orthogonal polarizations are employedin a switching scheme, and a tag is aligned diagonally between those twopolarizations, the polarization power loss is theoretically identical tothat associated with a link between circular and linear polarizedantennas. A second disadvantage is that the reader must send switchcommands to control the antennas. The embodiments described and taughtherein, in which a single antenna provides multiple linear (orelliptical with high eccentricity) polarizations, progressing from onelinear polarization toward another orthogonal linear polarization as afunction of increasing channel frequency, might not be practical for avast majority of links in RF and wireless communication links, and hencethese embodiments would not be readily apparent to antenna andcommunication systems engineers. For example, whereas orthogonalpolarizations, each associated with a discrete bandwidth that isisolated from the other by a guard band, are not uncommon to promotechannel isolation (e.g., in forward and return links), devising acommunication scheme with a multitude of closely spaced channels inorder to promote polarization diversity would be consideredprohibitively complex and expensive. Frequency hopping spread spectrumtechniques applied in other communication practices (i.e., non-RFID),could not readily adopt the single feed multi-linear polarizationsolution because links with requirements for a high degree ofdeterminism could not tolerate the frequent polarization misalignmentsthat would result; i.e., the polarization hopping would have to besynchronized, which would require a priori knowledge of the antennaorientations on each side of the link, thus rendering the solutionentirely impractical. The impracticality of the solution is furtherhighlighted by the consideration that the source would be transmittingenergy over a variety of polarizations, only one of which would reallybe optimal for the receiving end. In contrast, an RFID reader commonlyserves a multitude of tag clients exhibiting the full range ofpolarizations, such that power transmitted at any specific polarizationis not generally wasted. Furthermore, in systems in which antenna sizeand complexity are not extreme design considerations, circularlypolarized antennas are the natural choice to deal with uncertainty ofthe relative orientations between two antennas. The confluence of (i)RFID systems with a reader addressing up to a multitude of linearlypolarized tags of arbitrary orientation; (ii) RFID applications that aretolerable of non-determinism; and (iii) the incorporation of a frequencyhopping scheme for the purposes of interference mitigation andinteroperability present a problem space that is somewhat uniquerelative to historic RF and wireless communication scenarios. Hence,antennas specialized for this problem space have not been heretoforeconsidered. Said problem space is well addressed by a single feedmulti-linear polarization antenna, as taught herein, that progressesfrom one linear polarization toward another orthogonal linearpolarization as a function of increasing channel frequency.

Certain embodiments described above with reference to FIGS. 11A, 11B,and 20-23 may also be described as follows. A system may include an RFIDinterrogator configured for generating an RFID signal, where the channelfrequency of the RFID signal changes over time within a givenoperational bandwidth; at least one single feed patch antenna; and atleast one single feed line configured for feeding the RFID signal to itscorresponding single feed patch antenna at a single feed point, wherethe at least one single feed patch antenna is configured to transmit anelectromagnetic wave in response to and at the channel frequency of theRFID signal fed thereto from its corresponding single feed line. The atleast one single feed patch antenna is further configured such that theelectromagnetic wave exhibits (1) a polarization tilt angle that variesdepending on the channel frequency of the RFID signal, (2) asubstantially linear polarization at all channel frequencies of the RFIDsignal within the given operational bandwidth, and (3) a range ofpolarization tilt angles across the given operational bandwidth thatspans at least 70 degrees within a single quadrant. In some embodiments,the variation in the polarization tilt angle between a first channelfrequency of the RFID signal and a second, consecutive channel frequencyof the RFID signal does not exceed 45 degrees. In some embodiments, theelectromagnetic wave is characterized by a polarization axial ratiogreater than or equal to 3.0 dB, at all channel frequencies of the RFIDsignal within the given operational bandwidth. As in other embodimentsherein, the channel frequency of the RFID signal generated by the RFIDinterrogator may change over time according to a frequency hoppingspread spectrum protocol. Further and more specifically as to thevariation of polarization, the single feed patch antenna may beconfigured such that the electromagnetic wave has a first substantiallylinear polarization with a first polarization tilt angle in response tothe RFID signal when the RFID signal has a channel frequency at or neara lower end of the given operational bandwidth, and such that theelectromagnetic wave has a second substantially linear polarization witha second polarization tilt angle in response to the RFID signal when theRFID signal has a channel frequency at or near a higher end of the givenoperational bandwidth, the first substantially linear polarization andthe second substantially linear polarization being substantiallyorthogonal to one another (e.g., substantially horizontal and verticalpolarizations, or vice versa). Orthogonal polarizations are ones thatdiffer from one another by 90 degrees; polarizations are considered“substantially orthogonal” to one another even if they deviate fromorthogonality by up to 20 degrees, e.g., two polarizations that differfrom one another by 70 degrees are considered “substantiallyorthogonal.” Further in this regard, the single feed patch antenna maybe configured such that the electromagnetic wave has a substantiallylinear polarization at all channel frequencies within the operationalbandwidth and has a polarization tilt angle that varies between thepolarization tilt angle of the first substantially linear polarizationand the polarization tilt angle of the second substantially linearpolarization as the channel frequency of the RFID signal increases fromthe lower end of the given operating bandwidth to the higher end of thegiven operating bandwidth. Further, the single feed patch antenna may becharacterized by a lower resonant mode center frequency and an upperresonant mode center frequency, wherein the absolute value of thedifference between the lower resonant mode center frequency and theupper resonant mode center frequency, divided by the upper resonant modefrequency, is greater than 1.8 times the 2:1 VSWR bandwidth and lessthan 3.5 times the 2:1 VSWR bandwidth. Further, the single feed patchantenna may be characterized by a 2:1 VSWR bandwidth that is equal to orgreater than 40% of the given operational bandwidth and equal to or lessthan 55% of the given operational bandwidth. Further, the single feedpatch antenna may be characterized by a lower resonant mode, an upperresonant mode, a lower resonant mode center frequency, and an upperresonant mode center frequency, wherein a value of sigma is greater thanor equal to 0.8 and less than or equal to 1.5, where sigma is theabsolute value of the difference between the lower resonant mode centerfrequency and the upper resonant mode center frequency, divided by thegiven operational bandwidth, and wherein each of a 2:1 VSWR bandwidth ofthe lower resonant mode and a 2:1 VSWR bandwidth of the upper resonantmode divided by sigma is greater than or equal to 0.26 and less than orequal to 0.54.

Further, the single feed patch antenna may be characterized by one ormore of the following: (1) the single feed patch antenna comprises alength and a width, the length being different in magnitude from thewidth, (2) the single feed patch antenna comprises a shape that issquare or rectangular and that has one or more truncated corners, (3)the single feed patch antenna comprises a conductive region, theconductive region including a cutout thereof, and (4) the single feedline feeds the RFID signal to the single feed patch antenna at a cornerof the single feed patch antenna.

Further, the single feed patch antenna may comprise anon-ground-plane-side metallization, and the non-ground-plane-sidemetallization may be characterized by a length, a width not equal to thelength, an area equal to the product of the length and the width, and aperturbation area comprising an addition to or subtraction from thearea, wherein the ratio of the perturbation area to the area (i.e., theratio Ds/S) is equal to (2 sigma Bo)/(2+sigma Bo) for a Type B design,and is equal to (sigma Bo)/(2+sigma Bo) for a Type A design, where Bo isthe fractional operational bandwidth of the single feed patch antenna,as noted above. In a Type B design, the perturbation area readilyrelates to a width W and a length L.

In other embodiments, the single feed patch antenna may comprise anon-ground-plane-side metallization, and the non-ground-plane-sidemetallization may be characterized by a length, a width equal to thelength, an area equal to the product of the length and the width, and aperturbation area comprising an addition to or subtraction from thearea. In a first set of these other embodiments, the perturbation areamay comprise an addition to or subtraction from the area in a diagonaldirection of the single feed patch antenna, the diagonal directioncomprising a direction diagonal relative to the length and the width,and the single feed line may be disposed at a center of the length or ata center of the width, at one side of the single feed patch antenna, andbe parallel to the length or width. More specifically, in some cases,the perturbation area may comprise a subtraction from the area, thesubtraction comprising a portion cutout from an interior of the area, orthe subtraction comprising truncation of one or more corners (alsoreferred to as one or more chamfered corners) of the area, while in somecases the perturbation area may comprise an addition to the area, theaddition comprising a tab extending from a corner of the area. Further,a ratio of the perturbation area to the area may equal (sigmaBo)/(2+sigma Bo), where Bo is a fractional operational bandwidth of thesingle feed patch antenna. Still further, the single feed patch antennamay be characterized by a fractional 2:1 VSWR bandwidth greater than0.45 times the given operational bandwidth and less than 0.6 times thegiven operational bandwidth.

In a second set of these other embodiments (i.e., where the single feedpatch antenna comprises a non-ground-plane-side metallization, and thenon-ground-plane-side metallization is characterized by a length, awidth equal to the length, an area equal to the product of the lengthand the width, and a perturbation area comprising an addition to orsubtraction from the area), the perturbation area may comprise anaddition to or subtraction from the area in a direction parallel to thelength or the width of the single feed patch antenna, and the singlefeed line may be disposed at a corner the single feed patch antenna.More specifically, in some cases, the perturbation area may comprise asubtraction from the area, the subtraction comprising a portion cutoutfrom an interior of the area, or the subtraction comprising a portioncutout along a side of the area, while in some cases the perturbationarea may comprise an addition to the area, the addition comprising a tabextending from a side of the area, or the addition comprising anaddition extending along an entire side of the area (recalling that thepatch antenna is rectangular, it will be appreciated that where theaddition is along an entire side, say, the width, W, of the patch, thisaddition is equivalent to saying that the magnitude of the adjacentside, in this case, the length, L, is increased). Further, a ratio ofthe perturbation area to the area may equal (2sigma Bo)/(2+sigma Bo),where Bo is a fractional operational bandwidth of the single feed patchantenna. Still further, the single feed patch antenna may becharacterized by a fractional 2:1 VSWR bandwidth greater than 0.45 timesthe given operational bandwidth and less than 0.6 times the givenoperational bandwidth.

We now turn to a discussion of determining the orientation of an RFIDtag (and hence of the item attached to the tag) based on thepolarization of the RFID response signal received from the tag. Thisfeature can be implemented in FM RFID embodiments described herein. Howthis feature is performed is explained in the following example withreference to FIG. 12. An RFID reader 1205 has a reader antenna 1215 thattransmits an electromagnetic wave in either of two frequency bands andhaving linear polarization. The frequency bands (which may also bereferred to as channels) are a first band, ranging from frequency f₁through f₂, and a second band, ranging from frequency f₃ through f₄.Electromagnetic waves in the first band exhibit vertical polarization,A, and electromagnetic waves in the second band exhibit horizontalpolarization, B. The antenna of an RFID tag 1270 receives and respondsto the electromagnetic wave transmitted by the reader antenna 1215. Theresponsive electromagnetic wave sent by the RFID tag antenna 1270 has apolarization, C. The reader antenna 1215 receives the responsiveelectromagnetic wave sent by the RFID tag antenna 1270. If the receivedresponse is greater in power level or signal strength in the first band(f₁ through f₂) than in the second band (f₃ through f₄) (in other words,if the response signal received from the tag is predominantly within thefirst channel frequency), this result indicates that the polarization Cof the response is more closely aligned with the polarization A of thefirst band than with the polarization B of the second band, which may beexpressed as |{right arrow over (A)}·{right arrow over (C)}|>|{rightarrow over (B)}·{right arrow over (C)}|. If the received response isgreater in power level or signal strength in the second band (f₃ throughf₄) than in the first band (f₁ through f₂) (in other words, if theresponse signal received from the tag is predominantly within the secondchannel frequency), this result indicates that the polarization C of theresponse is more closely aligned with the polarization B of the secondband than with the polarization A of the first band, which may beexpressed as |{right arrow over (B)}·{right arrow over (C)}|>|{rightarrow over (A)}·{right arrow over (C)}|. The polarization C of theresponse corresponds with the orientation of tag antenna 1270 that sentthe response, and consequently with the orientation of the item (notshown) to which the tag (not shown) is attached. If the polarization Cof the response is more closely aligned with the polarization A of thefirst band, then the tag orientation is determined to be predominantlyvertical. If the polarization C of the response is more closely alignedwith the polarization B of the second band, then the tag orientation isdetermined to be predominantly horizontal. In general, of course, theorientation of the tag might lie somewhere in between, and weighting ormapping routines can be used to estimate the orientation of the tag. Thedetermination of the polarization of the response signal and theconsequent orientation of the tag antenna/tag may be made in conjunctionwith a processor (not shown) connected to or embedded within the reader1205.

We now turn to a discussion of FM RFID employing a frequency selectivesurface (FSS) in the propagation environment of the RFID signals, as anadditional element in an FM RFID system or method. The FSS may be usedto alter or increase scattering of the RFID signals (electromagneticwaves) sent by reader and/or tag, so as to increase field strength, orto eliminate regions of weak fields or nulls due to interference or thelike, or to increase the maximum received signal strength at the reader,as the RFID channel frequency varies. The FSS may also be used to alterthe polarization of the RFID signals as a function of the RFID channelfrequency. This feature can be implemented in FM RFID embodimentsdescribed herein. This feature is described now with reference to FIGS.13-17.

As throughout this disclosure, the FM RFID system represented in FIG. 13may employ an FHSS system. FIG. 13 shows an arrangement in which areader antenna 1315 (reader not shown in FIG. 13 but shown generally asreader 1205 in FIG. 12) is transmitting through an aperture 1314 in awall 1380 in a portal application. The region between the two walls1380, 1381 is a portal region (or portal) 1385 through which taggeditems (not shown) pass. Although such portals are sometimes establishedbetween reader antennas in open space on one or two sides of a portalregion, it is also common to erect conductive portal walls with theintent of better confining the electromagnetic radiation to promoteregions of higher field intensity to increase RFID signal penetrationinto collections of tagged items. Although this practice can inducestanding waves, and hence regions of higher field intensity, it can alsoconcurrently produce null regions.

Walls 1380, 1381 may be referred to as boundary walls. Although twowalls 1380, 1381 are illustrated, it is possible to employ only a singlewall (either the wall with the reader antenna or a wall without thereader antenna) or more than two walls. In this arrangement, FSSs 1382,1383 are applied on walls 1380, 1381, respectively, although it ispossible to apply an FSS on only a single wall (either the wall with thereader antenna or a wall without the reader antenna). While FSSs 1382,1383 are in this embodiment applied on the entirety of the respectivewalls 1380, 1381, it is possible in other embodiments to apply the FSSs1382, 1383 on only portions of the respective walls 1380, 1381. The FSSs1382, 1383 are frequency selective such that they present differentelectromagnetic boundary conditions, respectively, at differentfrequencies within the operating band. For example, the FSSs 1382, 1383may be configured such that at some channels within the FHSS band, thewalls 1380, 1381 appear as a perfect electrical conductor (PEC), whileat other of those channels the walls 1380, 1381 appear as an opencircuit boundary, or perfect magnetic conductor (PMC). Due to thedifferent boundary conditions at respective different frequencies withinthe operating band, the peaks and nulls vary throughout the portalregion 1385 so that fewer tagged items are missed. This situation isillustrated schematically in FIG. 14. FIG. 14 shows respectivecross-sections (showing modal peaks and nulls) of electric fieldamplitude 1465, 1466 between FSS walls 1480, 1481 of a portal region1485, at frequencies at which FSS walls 1480, 1481 present PEC boundaryconditions (solid line) and at frequencies at which FSS walls 1480, 1481present PMC boundary conditions (dashed line).

An FSS as described in this embodiment may scatter an electromagneticwave according to a first pattern (or alter the polarization of anelectromagnetic wave in a first manner) when the electromagnetic wavehas a frequency falling within a first bandwidth and scatter anelectromagnetic wave according to a second pattern (or alter thepolarization of an electromagnetic wave in a second manner) when theelectromagnetic wave has a frequency falling within a second bandwidth,the first and second patterns (and the first and second manners) beingdifferent, and the first and second bandwidths encompassing respectivelydifferent, non-overlapping ranges of frequencies. In this regard, thebandwidths may but need not be continuous, for example, the firstbandwidth may comprise two bandwidth segments on either side of thesecond bandwidth. In addition the pattern of scattering or manner ofalteration of polarization may change gradually (not abruptly) from thefirst pattern or manner to the second pattern or manner as the channelfrequency changes from a frequency within the first bandwidth to afrequency within the second bandwidth. The aforementioned configurationin which the first bandwidth comprises two bandwidth segments on eitherside of the second bandwidth is effectively a configuration having threebandwidth segments (three bands). It will be appreciated that any numberof bands may be employed in the FSS embodiments described here; thenumber of bands is not limited to two or three. The aforementionedalteration of polarization of an electromagnetic wave in a given mannermay be deemed a type or subset of the aforementioned scattering of anelectromagnetic wave according to a given pattern. Accordingly, in thisdiscussion of FSS embodiments, the term “pattern” will be understood toencompass both (i) a pattern defined by the angular distribution of theintensity of an electromagnetic wave and (ii) a pattern defined by thepolarization of an electromagnetic wave as a function of angle ((ii) mayalso be referred to as a pattern of altered polarization of anelectromagnetic wave).

FIG. 15 shows one of many possible layer architectures for an FSS 1582.In this illustrated example, FSS 1582 includes a solid conductive layer(ground plane) 1573 at the bottom, an insulating layer 1574 in themiddle, and a printed conductive layer (or pattern) 1575 on top. In thecontext of the portal application (e.g., an arrangement such as thatshown in FIGS. 13 and 14), the terms “bottom,” “middle,” and “top” areused to mean that where FSS 1582 is applied to a wall (e.g., 1380, 1381,1480, 1481) of the portal (1385, 1485), the bottom layer 1573 contactsthe wall, the middle layer 1574 lies between the bottom layer 1573 andthe top layer 1575, and the top layer 1575 faces the (ambientenvironment of the) portal region. Thus, in this context the terms“bottom,” “middle,” and “top” may also be understood asinterior/innermost (i.e., facing the wall), middle andexterior/outermost (i.e., facing the ambient environment of the portalregion). The printed conductive pattern 1575, in combination with thesolid conductive layer 1573, which is separated from the printedconductive pattern 1575 by a distance h, resonates at a frequency f₀. Atthat resonant frequency, the FSS 1582 resembles a PMC such that areflected tangential electric field is predominantly in phase with anincident tangential electric field. This result is in contrast to a PECboundary condition at which the reflected electric field is 180 degreesout of phase with the incident tangential electric field such that thetotal tangential part of the electric field vanishes at the boundary. Itis recognized that no physical conductor is perfect but that practicallymany conductors can be treated as such. It is also recognized that PMCsexist only as abstractions of physical phenomena in whichelectromagnetic fields behave as they would in the vicinity of ahypothetical PMC. In another embodiment, the Ohmic losses of the FSS ator near resonance are sufficiently high that the FSS impedance moreclosely resembles an impedance load that is matched to free space than aPMC. Thus, at frequencies near the resonance of the FSS, the wall maysignificantly absorb the impinging waves so as to preclude interferenceeffects, whereas at frequencies removed from resonance, but still withinthe RFID operating bandwidth, the FSS appears as a PEC.

An FSS may be instantiated by use of a metamaterial. A metamaterial is amaterial that is not found in nature, but is artificially synthesized insuch a manner as to have properties not ordinarily found in naturalmaterials. Further discussion of metamaterials, FSSs, and theirapplications may be found in U.S. Pat. Nos. 9,208,362, 9,465,965 and9,652,646, entitled “Methods, Systems and Apparatuses for RadioFrequency Identification,” a family of related applications which havesome inventors in common with the instant application, and all of whichapplications are hereby incorporated herein by reference in theirentirety.

Although the concept of frequency multiplexed scattering was describedabove in a portal application (FIGS. 13 and 14), there are many othercontexts in which an FSS or metamaterial scattering object may be usedto preferentially scatter the FHSS RFID signal transmitted by, orreceived by, the RFID reader antenna. Similarly, the FSS or metamaterialobject may be used to alter the polarization of the FHSS RFID signaltransmitted by, or received by, the reader antenna. FIG. 16 shows anarrangement for use with FM RFID, including an FSS 1682 in proximity toa distributed current sheet 1678. The current sheet 1678 may be any ofvarious antenna elements with bidirectional radiation, and the arrows inFIG. 16 represent electric current vectors established on the antenna.At some frequency bands within the FHSS spectrum, the FSS 1682 acts as aPEC. When the separation distance “d” between FSS 1682 and current sheet1678 is less than a quarter-wavelength, the mirror image (i.e.,reflected radiation) tends to cancel radiation in the direction normalto the current sheet 1678 and accentuate radiation at oblique angles. Atother frequency bands within the FHSS spectrum, the FSS 1682 acts as aPMC that (given the same condition regarding separating distance d)accentuates the radiation intensity in the direction normal to thecurrent sheet 1678 and diminish obliquely directed radiation. The FSS1682 may be constructed with layers such as those shown in FIG. 15.

As discussed above, an FSS may generally be employed with theembodiments disclosed herein. Accordingly, a basic FM RFID systememploying an FSS may be described as follows. Such a system may includean RFID interrogator configured for generating an RFID signal, where thechannel frequency of the RFID signal changes over time within anoperating bandwidth; at least one antenna; and a frequency selectivesurface. The operating bandwidth comprises a plurality of portionsthereof (which may be referred to as “bands”), e.g., a first band, asecond band, and a third band. The system may be arranged such that, forany two of these bands, the two bands do not overlap (or do notcompletely overlap) with each other (thus, the two bands encompassrespective ranges of frequencies that are mutually exclusive (or partlymutually exclusive, i.e., portions of their respective ranges aremutually exclusive)). According to the system, the RFID signal is to befed to the at least one antenna, and the at least one antenna isconfigured to transmit an electromagnetic wave in response to the RFIDsignal fed thereto. Further according to the system, the frequencyselective surface may comprise a surface whose electromagneticcharacteristics vary depending on a frequency of an electromagnetic waveimpinging thereon. The electromagnetic wave impinging thereon may be theelectromagnetic wave transmitted by the at least one antenna and/or anelectromagnetic wave transmitted by an RFID tag for reception by the atleast one antenna, whether in response to the electromagnetic wavetransmitted by the at least one antenna or not.

As mentioned above, the frequency selective surface may comprise asurface whose electromagnetic characteristics vary depending on afrequency of an electromagnetic wave impinging thereon. Specifically,the frequency selective surface may be configured to present as aboundary condition a surface impedance that changes according to thefrequency of an electromagnetic wave impinging thereon (e.g., accordingto the channel frequency of an RFID signal). The resulting pattern(i.e., of scattering or altered polarization of the electromagneticwave) produced by the frequency selective surface changes as a functionof the frequency of the wave, and these changes in pattern, over aplurality of frequencies (e.g., channel frequencies of the RFID signal),improve the odds that the reader will be able to communicate with alltags in the environment.

Continuing the description presented in the preceding two paragraphs,according to some embodiments, when the channel frequency of the RFIDsignal falls within the first portion of the operating bandwidth(“band”), a first surface impedance is established on the FSS thatalters the electromagnetic wave (impinging on the FSS) according to afirst pattern; when the channel frequency of the RFID signal fallswithin the second portion of the operating bandwidth (“band”), a secondsurface impedance is established on the FSS that alters theelectromagnetic wave (impinging on the FSS) according to a secondpattern; and when the channel frequency of the RFID signal falls withina third portion of the operating bandwidth (“band”), a third surfaceimpedance is established on the FSS that alters the electromagnetic wave(impinging on the FSS) according to a third pattern. In somearrangements, the first and third boundary conditions (surfaceimpedances), and hence the resulting electromagnetic wave patternsproduced thereby, are at least substantially the same; in otherarrangements, they are not. In general (although it is not required),the system is arranged such that at least one of the first, second, andthird surface impedances differs from the other two. Again, as noted,the resulting or altered electromagnetic wave patterns discussed heremay refer to scattering of the electromagnetic wave or altering of thepolarization of the electromagnetic wave. Again, the electromagneticwave (impinging on the FSS) discussed in this embodiment may be theelectromagnetic wave transmitted by the at least one antenna (of the FMRFID system employing the FSS) and/or an electromagnetic wavetransmitted by an RFID tag for reception by the at least one antenna,whether in response to the electromagnetic wave transmitted by the atleast one antenna or not. Also, as noted, the first, second and thirdportions of the operating bandwidth (“bands”) may not be completelyseparated, i.e., there may be some overlap between them. Consequently,there may be some variation of the resulting electromagnetic wavepatterns even within one of these three portions of the operatingbandwidth. In other arrangements, there may be only two bands, suchthat, when the channel frequency of the RFID signal falls within theband, a first surface impedance is established on the FSS that altersthe electromagnetic wave (impinging on the FSS) according to a firstpattern, and when the channel frequency of the RFID signal falls withinthe second band, a second surface impedance is established on the FSSthat alters the electromagnetic wave (impinging on the FSS) according toa second pattern. In other arrangements, there may be more than threebands.

Continuing the description presented in the preceding three paragraphs,as non-limiting examples, three common surface impedance boundaryconditions are: perfect electrical conductor (short circuit), perfectmagnetic conductor (open circuit), and resistive (matched termination).In a non-limiting exemplary arrangement, the second band comprises arange of frequencies that is generally between the range of frequenciesof the first band and the range of frequencies of the third band, thoughsome overlap of adjacent bands may exist. (Thus, the second band may bereferred to as the “center” band.) In this arrangement, over the firstand third bands, the FSS may present as a boundary condition a surfaceimpedance at least substantially equivalent to a short circuit, and overthe second band the FSS may present as a boundary condition a surfaceimpedance at least substantially equivalent to an open circuit.Alternatively, the FSS may present at least substantially a shortcircuit over the first and third bands and a matched resistivetermination over the second band. In both of these cases, the FSSresonates in the second (center) band. In an alternate arrangement theFSS resonates near one edge of the operating bandwidth, and there areonly two (rather than three) distinct bands. Another application of FSS,or an analogue thereof, in the context of FM RFID, similar to the portalapplication (FIGS. 13 and 14), is a smart shelf application. In such asmart shelf application, one or more shelves are employed, and one ormore surfaces of the shelf(ves) include one or more embedded waveguideseach with one or more frequency dependent loads (described below). Afrequency dependent load is the zero-dimensional, or point, analogue ofan FSS. Similarly as described with respect to the portal, the frequencydependent loads present different boundary conditions as a function offrequency within the FHSS band of operation. Yet another application ofFSS or frequency dependent loads, in the context of FM RFID, is a cavityor waveguide application. In this application, one or more cavity orwaveguide walls have an FSS, metamaterial surface, or frequencydependent load(s) such that the surface impedance varies as a functionof frequency within the FHSS operating spectrum, or a waveguide isterminated by a frequency dependent load. In this arrangement, a feed orantenna radiates into, or launches onto, the cavity or waveguide inorder to read RFID-tagged items contained within.

FIG. 17 shows a smart shelf including waveguides and frequency dependentloads, which are examples of reduced dimensionality analogs to thepreceding FSS or metamaterial arrangements. It has been shown thatalternating open and short terminations to smart shelf transmissionlines or surface waveguides can increase the communication capability byreducing the effect of nulls in standing waves (see, e.g., “A PlanarWaveguide Sheet with Switched Open/Short Termination for Smart-ShelfSystem”, K. Chen et al., Japan, IEEE 2012). In this reference, theauthors describe switching the load between an open circuit and a shortcircuit. The resultant effect is to eliminate null regions similarly asdescribed herein using an FSS or metamaterial. However, FIG. 17illustrates an arrangement that may accomplish similar effects withoutthe complexity associated with additional switches and controlcircuitry. More specifically, such similar effects are achieved usingfrequency dependent loads, Z_(L), as shown in the smart shelf 1701example of FIG. 17. The load Z_(L) may be a lumped element resonatorsuch that it appears as an open circuit over one or more first portionsof the FHSS operating spectrum, and as a short circuit over one or moresecond portions of the FHSS operating spectrum. In the arrangement ofFIG. 17, lines 1715 may be electromagnetic transmissive elements, suchas open (at least partly open) transmission lines, microstrip lines,coplanar waveguides, or the like. The ends of lines 1715 opposite theload Z_(L) ends may interface with an interrogator (not shown in FIG. 17but shown generally as reader 1205 in FIG. 12) and may be referred to asreader (interrogator) interface ports. Alternatively, these ends mayinterface with one or more antennas (not shown in FIG. 17 but showngenerally as antenna 1215 in FIG. 12) that receive signals from, ortransmit signals to, an interrogator (not shown). Each line 1715 carriesan RFID signal between the respective reader interface port (or antenna)and the respective load Z_(L). As stated, each of the load ends Z_(L)varies in its behavior or presents different electromagneticcharacteristics to the signal transmitted down the respective line 1715from the interrogator (or antenna) interface end, as a function of thefrequency of that signal (e.g., the channel frequency of an RFIDsignal). The number of lines 1715, each terminating in a load Z_(L), mayvary from the number illustrated in FIG. 17. Smart shelf 701 may be anopen structure (i.e., not enclosed, open to the environment), or it mayreside on a face of a fully or partially enclosed structure. Thus, atleast one of lines 1715 may be mounted on an open surface. As anon-limiting alternative, at least one of lines 1715 may be mounted on asurface of a wall of a cavity or waveguide, the cavity or waveguidecoupling the RFID interrogator and the at least one line 1715.

The following discussion provides non-limiting examples of more detailedelaborations of smart shelf 1701.

In these examples, the smart shelf 1701 may have four lines 1715 ratherthan three as shown in FIG. 17. Lines 1715 may be at least partly opentransmission lines, each having characteristic impedance Zo. Each loadZ_(L) may be implemented as lumped element, distributed elements, or acombination of lumped and distributed elements. In a first example, eachload Z_(L) may be implemented in the form of a 2nd-order parallelcircuit with symbolic lumped elements, e.g., a resistor, a capacitor andan inductor in parallel, sharing a common node or junction point withrespective line 1715, and with an electrically short distance betweeneach pair of adjacent ones of these three elements (R, C, L). Ingeneral, however, each load Z_(L) may comprise a parallel circuit, aseries circuit, or a combination of series and parallel circuits.Continuing with the first example (where load Z_(L) is implemented inthe form of a 2nd-order parallel circuit with symbolic lumped elementsas described above), each load Z_(L) may be nominally identical, e.g.,each resistor may have resistance R₁, each capacitor may havecapacitance C₁ and each inductor may have inductance L₁. Continuing withthe first example, the length of each transmission line 1715 may bemλ/4, where m is an odd or even integer. More generally, depending onthe frequency of the impinging signal, the design of load Z_(L), and thelength of transmission line 1715, the load Z_(L) impedance may appear asan open circuit, a short circuit, a matched circuit, or a generalimpedance at the reader or antenna interface.

In some instances, the load Z_(L) may appear at some frequencies as anopen or short circuit; in such cases, the at least partly opentransmission line 1715 is likely to radiate more than when the loadZ_(L) impedance is matched to that of the transmission line 1715. Whenthe transmission line 1715 radiates, it is likely to pick up tags thatare further removed from the transmission line 1715. At otherfrequencies, the load Z_(L) appears matched, and the transmission line1715 radiates much less, or essentially not at all. In this case, onlytags that are close to the transmission line 1715 are read. In thisfashion, based on the frequencies at which tags are read, and the phaseand signal strength of the return signal from the tag, a processorconnected to or internal to the reader may infer the tag's locationrelative to the transmission line 1715. In an embodiment, the resistanceR₁ equals the line characteristic impedance Zo.

Other design factors may be used to control the amount of radiation fromthe at least partly open transmission lines 1715. For example, thinnersubstrates typically result in less radiation. According to a secondexample, the multiple reader interface ports may be connected to adistributed diplexer (not shown in FIG. 17 but shown generally asdiplexer 610 in FIG. 6) and, assuming the transmission line 1715 lengthsare suitably controlled, all of the transmission lines 1715 may becollectively interfaced to a single reader port (as in FIG. 6 alltransmission lines 615 are collectively interfaced to a single readerport 606). In this second example, the diplexer controls the linedistances between adjacent lines 1715 and between the reader port andthe closest lines 1715 (these line distances are analogous to theinterface spacings m₁L, m₂L, m₃L between adjacent lines 615 and theinterface spacings n₁L, n₂L between the reader port 606 and the closestlines 615 in FIG. 6). Continuing with the second example, each loadZ_(L) may be the same 2nd-order parallel circuit with symbolic lumpedelements as for the previous example, except that in this example thevalues of the elements may all be different, e.g., the respectiveresistors may have resistances R₁, R₂, R₃ and R₄, the respectivecapacitors may have capacitances C₁, C₂, C₃ and C₄, and the respectiveinductors may have inductances L₁, L₂, L₃ and L₄. Continuing with thesecond example, each frequency dependent load Z_(L) has a differentresonant frequency, and the length of each transmission line 1715 (i.e.,from load Z_(L) to the diplexer line leading to the adjacenttransmission line 1715) is controlled to be an odd integer multiple of aquarter wavelength (mλ/4, where m is an odd integer) such that thestructure directs most of the power to the particular load Z_(L) that isresonant at the current channel frequency. At the resonant frequency,the load Z_(L) impedance is equal to the resistance shown in therespective parallel circuit. Far below or above the resonant frequency,the load Z_(L) appears as a short circuit.

A third example is the same as the second example, except that theresistor, capacitor and inductor comprising the load Z_(L) at the end ofeach transmission line 1715 are connected in series rather thanparallel. Again, any of these elements could be realized by distributedcomponents rather than lumped elements. As with the case of parallelloads, the load Z_(L) impedance at resonance is equal to the respectiveresistance shown. However, far below or above resonance, the load Z_(L)impedance tends toward an open circuit. In this case, the transmissionline 1715 is constrained to be an even integer multiple of a quarterwavelength (nλ/4, where n is an even integer). In addition to the first,second and third examples, smart shelf 1701 may include one or moretransmission lines 1715 having series loads and one or more transmissionlines 1715 having parallel loads, rather than all transmission lines1715 having loads of the same kind (parallel or series).

As described herein, a smart shelf or smart surface, such as describedabove with reference to FIG. 17, may be characterized as the followingsystem. The system may include an RFID interrogator configured forgenerating an RFID signal, wherein a channel frequency of the RFIDsignal changes over time within an operating bandwidth. The system mayfurther include one or more electromagnetic transmissive elements eachextending between a first end thereof and a second end thereof, each ofthe electromagnetic transmissive elements electrically coupled with theRFID interrogator at the first end thereof, each of the electromagnetictransmissive elements comprising a frequency dependent load at thesecond end thereof and configured for transmitting the RFID signal fromthe RFID interrogator to the frequency dependent load, wherein thefrequency dependent load presents different electromagnetic impedancecharacteristics to the RFID signal transmitted to the frequencydependent load depending on the channel frequency of the RFID signal. Insome embodiments, the electrical coupling of the electromagnetictransmissive elements with the RFID interrogator is established via anantenna (or a plurality of antennas). That is, the interrogator radiatesto a shelf antenna, which is connected to one or more of theelectromagnetic transmissive elements, at the first end(s) thereof. Theshelf antenna receives the signal from the interrogator and transmits itto the electromagnetic transmissive element(s).

While the discussion heretofore has described to one of ordinary skillin the art various methods of FM RFID using systems disclosed herein,nonetheless a limited and non-exhaustive group of exemplary methods orportions of such methods will now be described even more explicitlybelow.

FIG. 18 is a flow chart illustrating a method of FM RFID 1800. Method1800 illustrates the aforementioned first manner of operation offrequency multiplexing, in which signals are selectively distributed todifferent antennas/feed lines/transmission lines/etc., depending on thechannel frequency of the signals. Method 1800 may be employed with asystem including a plurality of antennas, a plurality of antenna feedlines (even if only a single antenna is included), or a plurality of(e.g., at least partially open) transmission lines. Method 1800 will bedescribed in two different ways: first, as a linear series of steps and,second, as an iterative loop. At step 1805, an RFID signal is generatedat one of a plurality of channel frequencies within an operatingbandwidth. An RFID interrogator may perform this step. As explainedbelow in the iterative loop description of method 1800, step 1805 isperformed multiple times. Each time, the RFID signal may be generated ata different one of the plurality of channel frequencies; theinterrogator may cause the channel frequency of the RFID signal tochange over time, e.g., according to an FHSS protocol. At step 1810, thegenerated RFID signals are distributed (or fed or transmitted) to aplurality of electromagnetic transmissive elements, respectively, atdifferent times, depending on the channel frequency of the RFID signal.In addition, the magnitude, or weighting, and the phase of the RFIDsignal generated and distributed to a respective electromagnetictransmissive element varies over time, depending on the channelfrequency of the RFID signal. The plurality of electromagnetictransmissive elements may be a plurality of antennas, a plurality ofantenna feed lines, or a plurality of (e.g., open) transmission lines.The antennas may but need not be narrow band antennas including a firstnarrow band antenna and a second narrow band antenna, the first narrowband antenna configured to filter out frequencies falling outside of afirst range of frequencies and the second narrow band antenna configuredto filter out frequencies falling outside of a second range offrequencies, the first range of frequencies being at least partiallydifferent from the second range of frequencies. Step 1810 may beperformed by a diplexer, which may include a filter of any of varioustypes, or by other means (e.g., narrowband antennas, adjustment oflengths of transmission lines/adjustment of impedance of electromagnetictransmissive element relative to impedance of transmission line fromwhich the generated RFID signal is distributed to the electromagnetictransmissive element), as described herein. At step 1815, a firstelectromagnetic signal having a first channel frequency is transmittedin response to a first distributed RFID signal. This step may beperformed by one of the plurality of electromagnetic transmissiveelements, to which the first RFID signal was distributed. At step 1820,a second electromagnetic signal having a second channel frequency istransmitted in response to a second distributed RFID signal. This stepmay be performed by one of the plurality of electromagnetic transmissiveelements, to which the second RFID signal was distributed. The firstchannel frequency may differ from the second channel frequency. In thiscase, the first and second electromagnetic signals may be transmitted bydifferent ones of the plurality of electromagnetic transmissiveelements, the first and second RFID signal having been distributed tothe different ones, respectively. The first and second channelfrequencies may be overlapping or not overlapping. The first and secondchannel frequencies may be close/adjacent or not close/adjacent. Thefirst electromagnetic signal may have a first polarization, and thesecond electromagnetic signal may have a second polarization, the firstpolarization and the second polarization being characterized bydifferent orientations. (In this case, method 1800 illustrates acombination of the aforementioned two manners of operation of frequencymultiplexing, namely, the first manner of operation, in which signalsare selectively distributed to different antennas/feedlines/transmission lines/etc., depending on the channel frequency of thesignals, and the second manner of operation, in which signals excitedifferent polarizations, depending on the channel frequency of thesignal.) The first polarization and the second polarization may beorthogonal to each other, e.g., one may be horizontal and the other maybe vertical. Where the plurality of electromagnetic transmissiveelements is a plurality of at least partially open transmission lines,upon transmission of the aforementioned first and second electromagneticsignals, electromagnetic energy may be transmitted to an RFID tag bynear-field coupling or radiation from the first and secondelectromagnetic signals.

At step 1825, a third electromagnetic signal transmitted by an RFID tagmay be received. The third electromagnetic signal may have beentransmitted by the RFID tag in response to the transmitted first orsecond electromagnetic signal. As explained below in the iterative loopdescription of method 1800, step 1825 may be repeated. At step 1830,information pertaining to a location and/or orientation of the RFID tagmay be determined based on the third electromagnetic signal receivedfrom the RFID tag. This step may be performed by the RFID interrogatorand/or a processor/processor logic associated therewith. As per thediscussion above regarding the definition of the term “localization,” itshould be understood that this step may involve determination oflocation, position, distance (e.g., between tag and interrogator),presence, existence, movement (e.g., speed, direction), etc., as well astracking of an RFID tag. With regard to orientation/bearing, step 1830may involve determining an orientation of an RFID tag based on thechannel frequency of the third electromagnetic signal sent by the RFIDtag in response to the first or second electromagnetic signal, asdescribed above with reference to FIG. 12. Specifically, if the thirdelectromagnetic signal is predominantly within the first channelfrequency, the tag is determined to have an orientation corresponding tothe first polarization, and if the third electromagnetic signal ispredominantly within the second channel frequency, the tag is determinedto have an orientation corresponding to the second polarization (thisconclusion assumes the first electromagnetic signal has the firstpolarization and the second electromagnetic signal has the secondpolarization as described above). While the discussion here has referredto determination of location, etc. of an RFID tag, it should beunderstood that such determination applies also, by extension, to theitem to which the tag is attached. As explained below in the iterativeloop description of method 1800, step 1830 may be repeated. In repeatedinstances of step 1830, determination of location, etc. may be refinedrather than newly established. At optional step 1835, which may beperformed prior to step 1815, a frequency selective surface (FSS) isprovided. The FSS may be disposed within a (spatial) range of thetransmitted first electromagnetic signal (wave) and/or a (spatial) rangeof the transmitted second electromagnetic signal (wave) (described insteps 1815 and 1820 above). Assuming step 1835 has been so performed,thereafter at optional step 1840, the transmitted first electromagneticsignal (wave), the transmitted second electromagnetic signal (wave),and/or a third electromagnetic signal (wave) sent by an RFID tag (e.g.,in response to the transmitted first or second electromagnetic signal(wave)) is scattered and/or the polarization of any of these signals(waves) is altered. The scattering is according to a particular pattern,the particular pattern depending on the channel frequency of therespective signal (wave). The altering of the polarization is accordingto a particular manner, the particular manner depending on the channelfrequency of the respective signal (wave). Step 1840 may be performed bythe FSS. In step 1835, where at least partially open transmission linesare used rather than antennas, one or more frequency dependent loads maybe provided instead of the frequency selective surface. In this case,each of the at least partially open transmission lines may extendbetween a respective first end thereof, to which the RFID signal is fedfrom an RFID interrogator (optionally via an antenna, as describedabove), and a respective second end thereof, opposite the first end. Thefrequency dependent load is provided at the second end of each of one ormore of the at least partially open transmission lines. The frequencydependent load presents different electromagnetic characteristics (e.g.,surface impedance boundary conditions, as described above) to a signalencountering the frequency dependent load, depending on a channelfrequency of the signal. Use of a frequency selective surface orfrequency dependent load may assist in the determination operation step(1830) as described in this disclosure.

Method 1800 will be now elaborated on, in order to describe it as aniterative loop, which may be a more intuitive presentation. As mentionedabove, step 1805 may be repeated, and with each iteration of step 1805,the RFID signal may be generated at a different one of the plurality ofchannel frequencies. Thus, the channel frequency of the RFID signal maybe said to change over time, e.g., according to an FHSS protocol. As forstep 1810, its description given above in the linear-series-of-stepsdescription of method 1800 already contains within it the repetition oriteration that occurs. That is, in a first iteration of step 1810, anRFID signal generated at a first channel frequency is distributed at afirst time to a first one of the plurality of electromagnetictransmissive elements, depending on the channel frequency of the RFIDsignal. Following the first iteration of step 1810, step 1815 isperformed, in which a first electromagnetic signal having the firstchannel frequency is transmitted in response to the first distributedRFID signal (the RFID signal distributed in the first iteration of step1810). Following step 1815, a first iteration of step 1825 is performed.(Note that step 1820 is not performed at this time, because it is asecond iteration of the operation of step 1815; third and subsequentiterations of this operation may be performed, but they were notdescribed in the linear-series-of-steps description given above.)Following the first iteration of step 1825, a first iteration of step1830 is performed. At this point, the iterative looping occurs, andfollowing the first iteration of step 1830, the method returns to step1805, and repeats. In the second iteration of the method, at step 1805,the RFID signal may be generated at a second channel frequency, which isat least partly different from the first channel frequency of the RFIDsignal generated at step 1805 in the first iteration of the method. (Dueto the pseudo random nature of FHSS, it is possible that the RFID signalgenerated at step 1805 in a given iteration of the method may fallwithin the same channel frequency as the RFID signal generated at step1805 in a temporally adjacent iteration of the method.) Continuing withthe second iteration of the method, at step 1810, the RFID signalgenerated at the second channel frequency is distributed at a secondtime to a second one of the plurality of electromagnetic transmissiveelements, based on the channel frequency of the RFID signal (the secondelectromagnetic transmissive element is different than the firstelectromagnetic transmissive element, because the second channelfrequency is different than the first channel frequency). Following thesecond iteration of step 1810, step 1820 is performed: a secondelectromagnetic signal having the second channel frequency istransmitted in response to the second distributed RFID signal (the RFIDsignal distributed in the second iteration of step 1810). After step1820, a second iteration of step 1825, and subsequently a seconditeration of step 1830, are performed. As noted, in the second orsubsequent iterations of step 1830 a previously determined location,etc. may be refined, based on the electromagnetic wave transmitted bythe RFID tag and received in preceding step 1825, rather than a newdetermination made. In particular, where the respective electromagneticwaves transmitted by the RFID tag and received in different iterationsof step 1825 fall within different channel frequencies, the receivedwaves may be particularly useful in refining the determination oflocation, etc. Following step 1830, the iterative loop recurs. In thenext iteration, of course, step 1820 would be replaced by acorresponding step (namely, transmitting a third electromagnetic signalhaving a third channel frequency in response to a third distributed RFIDsignal) that was not described in the linear-series-of-steps descriptionof the method given above, as that description covers only the basic(and broadest) iterative case, that is, the case having only twoiterations. Finally, it should be noted that the iterative loop methodmay of course include optional steps 1835 and 1840. In that regard, step1835 need only be performed one time, e.g., during the first iterationof the method, and step 1840 may be repeated over successive iterations.

FIG. 19 is a flow chart illustrating another method of FM RFID 1900.Method 1900 illustrates the aforementioned second manner of operation offrequency multiplexing, in which signals excite different polarizations,depending on the channel frequency of the signal. Method 1900 may beemployed with a system including a single feed microstrip antenna withan associated single antenna feed line. It may also be employed by asystem having multiple antennas and/or feed lines. At step 1905, an RFIDsignal is generated. An RFID interrogator may perform this step. Thechannel frequency of the RFID signal may change over time within abandwidth. In another step (not illustrated), the channel frequency ofthe RFID signal may change over time according to an FHSS protocol. Atstep 1910, the RFID signal is fed to a microstrip antenna. Themicrostrip antenna may be a patch antenna. This step may be performed byan antenna feed line. At step 1915, an electromagnetic wave istransmitted in response to the RFID signal fed to the microstripantenna. The transmitted electromagnetic wave has a polarization thatvaries depending on the channel frequency of the RFID signal fed to themicrostrip antenna. This variation of polarization depending on thechannel frequency of the RFID signal has been described elsewhere in theinstant disclosure. Step 1915 may be performed by the microstripantenna.

Step 1915 may be performed as follows. An electromagnetic wave having afirst polarization may be transmitted by the microstrip antenna if thechannel frequency of the RFID signal falls within the first portion ofthe bandwidth, and an electromagnetic wave having a second polarizationmay be transmitted by the microstrip antenna if the channel frequency ofthe RFID signal falls within the second portion of the bandwidth, thefirst and second polarizations being orthogonal to one another.

Steps 1910 and 1915 may be performed as follows. The RFID signal may befed to a first microstrip antenna if the channel frequency of the RFIDsignal falls within a first portion of the bandwidth, and the RFIDsignal may be fed to a second microstrip antenna if the channelfrequency of the RFID signal falls within a second portion of thebandwidth, the first and second portions of the bandwidth comprisingrespectively different, but possibly overlapping, ranges of frequencies(step 1910). An electromagnetic wave having a first polarization may betransmitted by the first microstrip antenna if the channel frequency ofthe RFID signal falls within the first portion of the bandwidth, and anelectromagnetic wave having a second polarization may be transmitted bythe second microstrip antenna if the channel frequency of the RFIDsignal falls within the second portion of the bandwidth, the first andsecond polarizations being orthogonal to one another (step 1915).

Steps 1910 and 1915 may be performed as follows. The RFID signal may befed to the microstrip antenna via a first transmission line if thechannel frequency of the RFID signal falls within a first portion of thebandwidth, and the RFID signal may be fed to the microstrip antenna viaa second transmission line if the channel frequency of the RFID signalfalls within a second portion of the bandwidth, the first and secondportions of the bandwidth comprising respectively different, butpossibly overlapping, ranges of frequencies (step 1910). Anelectromagnetic wave having a first polarization may be transmitted bythe microstrip antenna if the channel frequency of the RFID signal fallswithin the first portion of the bandwidth, and an electromagnetic wavehaving a second polarization may be transmitted by the microstripantenna if the channel frequency of the RFID signal falls within thesecond portion of the bandwidth, the first and second polarizationsbeing orthogonal to one another (step 1915).

The description of steps 1825, 1830, 1835 and 1840 of method 1800 isapplicable to method 1900, the changes being made in view of thedifferences between method 1800 and method 1900 as described heretofore(e.g., method 1900 may be performed by a system including at least onesingle feed microstrip antenna and a single antenna feed line, whilemethod 1800 may be performed by a system including a plurality ofantennas, feed lines, or transmission lines).

As with method 1800, so too analogously method 1900 may be performed asan iterative loop, as will be understood by one of ordinary skill nowhaving the benefit of the description of method 1800 given herein andthe remainder of this disclosure.

We now turn to a discussion of various implementationexamples/prototypes, test results, refinements and applications.

One prototype of an RFID-enabled pantry drawer for the InternationalSpace Station (ISS) includes four loop antennas in the drawer that areconnected to an RFID reader. Although the loop antennas are highlyeffective at exciting RF fields in the drawer, the standoff distancefrom each antenna to the drawer wall is about 0.8 inches. This distancerepresents a significant loss of storage volume and also presents anobtrusion on which articles can become snagged. Accordingly, in someinstances end users request that a planar cover for the antennas beplaced over the entire wall of the drawer. This cover, of course,represents an even larger loss of storage volume. This prototype wascompared with two frequency multiplexing prototypes, described below.

A second prototype is a four-element (four patch antennas),vertically-polarized feed comprising copper-clad Rogers 3010 board ofthickness 0.635 mm (“VP patch”). A third prototype is a similarfour-element (four patch antennas) circularly-polarized feed, usingelements of the type shown in FIG. 11 (“CP patch”). The second prototypewas tested with a series of full-wave simulated electric current on thefeed elements at four different frequencies, each frequencycorresponding to a center band of one of the elements. In the tests, oneof the four elements showed a stronger current excited on it compared tothe other three, thus indicating that the signal is largely multiplexedto that element. Simulation results of the power delivered to each ofthe four patches as a function of frequency showed four power signalsthat were somewhat overlapping but had distinct peaks, with higher peaksat the lowest and highest frequencies.

To compare performance, the three prototypes (loop, VP patch, and CPpatch) were alternately secured in the prototype drawer and connected toan EPCglobal Class 1 Generation 2 Impinj Speedway reader. The drawer wasfilled with 150 Styrofoam blocks, each block containing an Alien“Squiggle” RFID tag. The number of tags read and the total number ofreads were recorded for six different arrangements of the tags, the tagsbeing disposed in a different orientation in each arrangement.

The results were tabulated in terms of the percentage of tags read forthe loop, the VP patch, and the CP patch, for the six tag orientations.In most cases, the loop outperformed the multiplexed feeds. However, inall cases the multiplexed feeds read more than 90% of the 150 tags, andfor some orientations the multiplexed feeds outperformed the loop feed.These results are quite positive considering that the multiplexed feeds(VP patch, CP patch) are approximately 30 times thinner than the loopantenna, and microstrip antenna efficiency degrades as the thicknessdecreases.

A fourth prototype is described as follows. This prototype may bethought of as a variant or elaboration of system 700, and it providesboth spatial and polarization diversity for a smart drawer (orcontainer) designed for the International Space Station (ISS). On eachof two interior opposing sides of the drawer, a pair of thin panel(patch), frequency multiplexed antennas are provided, side by side. Eachpair includes a vertically polarized ¼-wave antenna and a horizontallypolarized ½-wave antenna. (This structure may be achieved by having the¼-wave antenna oriented vertically and the ½-wave antenna orientedhorizontally, the latter thus being rotated by 90 degrees relative tothe former, as described for some embodiments in the description of FIG.7 above.) The ¼-wave element was selected for vertical orientation dueto lack of space in the vertical dimension within the drawer; if thevertical dimension is sufficient, the pair may include a verticallypolarized ½-wave antenna and a horizontally polarized ¼-wave antenna.Note that each element is characterized by one mode and onepolarization. However, the polarizations on opposing sides of the boxare complementary, that is, the horizontally polarized antenna of thefirst pair (one side of the drawer) is opposed to the verticallypolarized antenna of the second pair (the other side of the drawer), andthe vertically polarized antenna of the first pair (one side of thedrawer) is opposed to the horizontally polarized antenna of the secondpair (the other side of the drawer). The attached reader uses two portsto connect to the two pairs of antennas, respectively, so that fourantenna elements are effectively fed with only two feeds. Alternatively,the two pairs of antennas may be fed via a single reader port.

In the following discussion a refinement is presented. Although certainembodiments described above incorporate the bandpass function inherentin narrow band antennas (e.g., printed patch antennas) to achievefrequency multiplexing, printed or “lumped element” filters mayalternatively be applied independent of any band-limiting function ofthe patch antenna. There are at least two advantages that may be derivedfrom this approach, notwithstanding the fact that in some instancesutilizing the patch's passband characteristics simplifies design andconstruction. One advantage relates to the fact that narrowing thebandwidth of the patch antenna increases its quality factor (Q), whichresults in a lower radiation efficiency. Segregating the patch radiationand the narrowband filtering function permits optimization of the patchas a radiator and optimization of the filter function so as to achieve ahigh Q for frequency multiplexing. In this arrangement, one option is toprint the filter elements external to the patch. A second option is toutilize surface acoustic wave (SAW) bandpass filters. SAW filters havefound extensive use in RF and microwave applications due to theextremely high Q that is achievable. For purposes of embodimentsdescribed herein, this feature may be used to refine spatiallocalization or extend the domain of localization over a larger areawhile retaining a fixed spatial localization accuracy.

In regard to this refinement, a SAW bandpass filter on a carrier withcoaxial RF (e.g., SubMiniature version A (SMA)) connectors on each sidewas prototyped and tested. The two-port S-parameter response of thefilter was measured using a vector network analyzer, and the resultsshow that the characteristic response is well-modeled by a parallel RLCtank circuit. The recorded two-port parameter set was imported into asimulation tool. A second two-port parameter set was created bytranslating the original filter frequency response. Simulations werethen conducted to determine how close, in spectrum, two such filterscould be placed in order to determine how many localization channelscould be created. The results showed that two such SAW filters can bespaced as closely as to have 2.5 MHz of center-band separation betweenthem without significantly affecting the passband of either filter. Tobe sure, even narrower bandwidths can be achieved with SAW devices.Although the insertion loss increases with narrower bandwidth, theselosses can be compensated for by increasing the transmit power. In somecases, however, lower transmit power is actually used. For example,cross-talk to adjacent smart shelves can be reduced through lowertransmit power applied to a shelf.

In the following discussion, examples of additional applications ofembodiments described herein are presented. One example applicationpertains to athletic training, where information pertaining to path orfoot placement of an athlete is sought. For example, if a baseballpitcher's stride lands on the pitcher's glove side of the center line tothe catcher, a rotational motion around the central body axis isestablished, making it more difficult to consistently deliver strikes.On the other hand, if the pitcher's stride lands on the other side ofthe center line, the pitcher tends to “throw across his body,” typicallyreducing the achievable speed of the pitch and inducing additionalstress on the throwing elbow. The length of the stride is alsoimportant, as reducing the distance from the release point to home platereduces the time afforded the batter, by increasing the so-called“virtual velocity” of the pitch. Although motion imagery (e.g., video)can capture such foot position, the use of FM RFID would provideautomated and real-time/near real-time evaluation as compared to videoanalysis that typically requires more human analysis and occurs muchlater. Footwork is likewise important for baseball players in otherpositions, such as catcher or shortstop. Gymnastics, fencing, tennis,football, and boxing are examples of other sports in which athletes maybenefit from real-time feedback regarding foot positioning such as maybe provided by FM RFID embodiments described herein.

Embodiments described herein can greatly facilitate such real-timeassessment of foot position and orientation. To do so, footwear can beequipped with RFID tags. RFID tags can be made lightweight and flexible,thus being relatively imperceptible to the wearer. The multiplexed feedscan be embedded under a carpet, artificial turf, exercise mat, or thenatural playing surface (e.g., Earth) of an athletic field.

Extension to a full football field would allow real-time tracking andrecording of player movements as plays unfold. However, given the sizeof the field and the domain over which a reader can operate, suchapplication may be quite expensive. Other applications, such as thetracking of gaining pieces (e.g., cards, chips, etc.) by FM RFID on asmart-grid table (e.g., in a casino), would be quite economical.

In light of the principles and exemplary embodiments described andillustrated herein, it will be recognized that the exemplary embodimentscan be modified in arrangement and detail without departing from suchprinciples. Also, the foregoing discussion has focused on particularembodiments, but other configurations are contemplated. In particular,even though expressions such as “in one embodiment,” “in anotherembodiment,” “in a version of the embodiment” or the like are usedherein, these phrases are meant to generally reference the range ofpossibilities of embodiments, and are not intended to limit thedisclosure to the particular embodiments and configurations describedherein. As used herein, these terms may reference the same or differentembodiments that are combinable into other embodiments.

Similarly, although exemplary processes have been described with regardto particular operations performed in a particular sequence, numerousmodifications could be applied to those processes to derive numerousalternative embodiments of the present disclosure. For example,alternative embodiments may include processes that use fewer than all ofthe disclosed operations, processes that use additional operations, andprocesses in which the individual operations disclosed herein arecombined, subdivided, rearranged, differently ordered or otherwisealtered.

In view of the wide variety of useful permutations that may be readilyderived from the exemplary embodiments described herein, this detaileddescription is intended to be illustrative only, and should not be takenas limiting the scope of the disclosure. What is claimed as thedisclosure, therefore, are all implementations that come within thescope of the following claims, and all equivalents to suchimplementations. In the claims, means-plus-function andstep-plus-function clauses are intended to cover the structures or actsdescribed herein as performing the recited function and not onlystructural equivalents, but also equivalent structures. Thus, while anail and a screw may not be structural equivalents in that a nailemploys a cylindrical surface to secure wooden parts together, whereas ascrew employs a helical surface, in the environment of fastening woodenparts, a nail and a screw may be equivalent structures.

What is claimed is:
 1. A system comprising: an RFID interrogatorconfigured for generating an RFID signal, wherein a channel frequency ofthe RFID signal changes over time according to a frequency hoppingspread spectrum protocol within an operating bandwidth, the operatingbandwidth comprising a first portion thereof and a second portionthereof, the first portion of the operating bandwidth not overlappingwith the second portion of the operating bandwidth; and one or moreelectromagnetic transmissive elements each extending between a first endthereof and a second end thereof, each of the electromagnetictransmissive elements electrically coupled with the RFID interrogator atthe first end thereof, each electromagnetic transmissive elementcomprising a passive frequency dependent load at the second end thereofand configured for transmitting the RFID signal from the RFIDinterrogator to the frequency dependent load, wherein each passivefrequency dependent load presents different electromagneticcharacteristics to the RFID signal transmitted thereto depending on thechannel frequency of the RFID signal, each passive frequency dependentload designed for appearing substantially as an open circuit or as ashort circuit at the second end for at least one channel frequency in atleast one of the first portion and the second portion of the operatingbandwidth such that, in operation, the system frequency multiplexes theRFID signal according to the channel frequencies arising from thefrequency hopping spread spectrum protocol.
 2. The system according toclaim 1, wherein at least one of the passive frequency dependent loadscomprises a lumped element resonator.
 3. The system of claim 1, whereinat least one of the passive frequency dependent loads comprises a narrowband antenna.
 4. The system according to claim 1, wherein each of theelectromagnetic transmissive elements is an at least partly opentransmission line having a length equal to an odd or even integermultiple of a quarter of a wavelength, the wavelength being a wavelengthat a center of the operating bandwidth.
 5. The system according to claim1, further comprising a distributed diplexer through which all of theelectromagnetic transmissive elements are jointly electrically coupledwith the RFID interrogator via a single port of the RFID interrogator.6. The system according to claim 1, wherein at least one of theelectromagnetic transmissive elements is mounted on an open surface. 7.The system according to claim 1, wherein at least one of theelectromagnetic transmissive elements is mounted on a surface of a wallof a cavity or waveguide, the cavity or waveguide coupling the RFIDinterrogator and the at least one electromagnetic transmissive element.8. A method, comprising: generating an RFID signal, wherein a channelfrequency of the RFID signal generated changes over time according to afrequency hopping spread spectrum protocol; frequency multiplexing theRFID signal to one or more electromagnetic transmissive elements atdifferent times, respectively, depending on the channel frequency of theRFID signal generated; transmitting a first electromagnetic signalhaving a first channel frequency in response to a first distributed RFIDsignal having the first channel frequency; and transmitting a secondelectromagnetic signal having a second channel frequency in response toa second distributed RFID signal having the second channel frequency,wherein the one or more electromagnetic transmissive elements comprisesa passive frequency dependent load.
 9. The method according to claim 8,wherein the RFID interrogator is compatible with an EPCGlobal RFIDprotocol and the frequency hopping spread spectrum protocol is definedthe EPCglobal protocol.
 10. The method according to claim 8, wherein thefrequency multiplexing of the RFID signal to each of the one or moreelectromagnetic transmissive elements, respectively, depending on thechannel frequency of the RFID signal generated, is performed using abandpass filter or a surface acoustic wave filter.
 11. The methodaccording to claim 8, wherein the frequency multiplexing of the RFIDsignal to each of the one or more electromagnetic transmissive elementsis performed by using a first narrow band antenna and a second narrowband antenna, the first narrow band antenna filtering out frequenciesfalling outside of a first range of frequencies and the second narrowband antenna filtering out frequencies falling outside of a second rangeof frequencies, the first range of frequencies being different from thesecond range of frequencies.
 12. The method according to claim 8,wherein the first electromagnetic signal having the first channelfrequency has a first polarization, and the second electromagneticsignal having the second channel frequency has a second polarization,the first polarization and the second polarization being characterizedby different orientations.
 13. The method according to claim 12, furthercomprising: determining an orientation of an RFID tag based on a channelfrequency of a third electromagnetic signal sent by the RFID tag inresponse to the first or second electromagnetic signal, wherein if thethird electromagnetic signal is predominantly within the first channelfrequency the tag is determined to have an orientation corresponding tothe first polarization, and if the third electromagnetic signal ispredominantly within the second channel frequency the tag is determinedto have an orientation corresponding to the second polarization.
 14. Themethod according to claim 8, further comprising: providing a frequencyselective surface within a range of the transmitted firstelectromagnetic signal having the first channel frequency and/or a rangeof the transmitted second electromagnetic signal having the secondchannel frequency; and scattering, in a particular pattern, and/oraltering a polarization, in a particular manner, of the transmittedfirst electromagnetic signal, the transmitted second electromagneticsignal, and/or a third electromagnetic signal sent by an RFID tag inresponse to the transmitted first or second electromagnetic signal, theparticular pattern and/or particular manner depending on a channelfrequency of the transmitted first electromagnetic signal, thetransmitted second electromagnetic signal, or the third electromagneticsignal, respectively.
 15. The method according to claim 8, furthercomprising receiving a third electromagnetic signal transmitted by anRFID tag in response to the transmitted first or second electromagneticsignal.
 16. The method according to claim 8, further comprisingdetermining information pertaining to a location and/or orientation ofan RFID tag based on a third electromagnetic signal received from theRFID tag in response to the transmitted first or second electromagneticsignal.
 17. The method according to claim 16, further comprisingrefining the determination of the information pertaining to a locationand/or orientation of an RFID tag based on another instance of receiptof a third electromagnetic signal from the RFID tag in response to thetransmitted first or second electromagnetic signal.
 18. The methodaccording to claim 8, further comprising transmitting electromagneticenergy to an RFID tag by near-field coupling or radiation from the firstand second electromagnetic signals, wherein the plurality ofelectromagnetic transmissive elements comprise the plurality of at leastpartially open transmission lines.
 19. The system of claim 1, whereinthe RFID interrogator is compatible with an EPCGlobal RFID protocol andthe frequency hopping spread spectrum protocol is defined by theEPCglobal protocol.